1 EE 136 PROJECT MASOOD NOOR DR. ZHOU 12/06/2003

2  INRUSH CONTROL INTRODUCTION: In “direct –off- line” switch mode supplies, minimizing the cost and size are the major consideration. In order to produce high-voltage DC supply for the converter, we have to use direct-off- line semiconductor bridge rectification with capacitive input filters. If the line inputs are switch directly to this type of rectifier capacitor arrangement, very large inrush currents will flow in the supply lines, input components, switches, rectifiers, and capacitors. Also, it may cause interference with other equipment sharing common supply line impedance. Different methods of “inrush current control” are used to reduce this stress. 1:SERIES RESISTORS: 2:THERMISTOR INRUSH LIMITING: 3:ACTIVE LIMITING CIRCUITS (TRIAC START CIRCUIT): SERIES RESISTORS: In this method a compromise must be made for low-power applications. The high value of resistance witch will give a low inrush current; will also be very dissipative under normal operation condition. Compromise selection must be made between acceptable inrush current and acceptable operating losses.

3  The series resistors must be selected to withstand the initial high voltage and high current stress (this accurse when the supply is first switched on). Special high current surge is suitable for this application. Also, rated wire wound types are often use too. Figure shows this operation. Two resistors should be used in positions R1 and R2. This has the advantage of effective parallel operation for low-voltage link and series operation for high-voltage link position. This is the inrush current at similar values for the two conditions, where single-rage input voltages are used, than a single inrush-limiting device may be fitted at position R3 at the input of the rectifiers.

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5  THERMISTOR INRUSH LIMITING:  Negative temperature coefficient thermostats (NTC) are often in the position of R1, R2, or R3 in low-power application. When the supply is first switch the resistance of the NTSC is high. This given then an advantage over normal resistors, this may be selected to give a low inrush current on initial switch-on and, since the resistance will fall when the thermostat self-heats under normal operating condition, excessive dissipation is avoided.  ACTIVE LIMITNG CIRCUITS (TRIAC START CIRCUIT):  The limiting device is better shorted out to reduce losses when the unit is fully operating for higher converters. Position R1 will normally be selected for the start resistor so that single triac or relay may be used. R1 can be shunted by a triac or relay after start-up. Since the start resistance can have a much higher value in this type of start up circuit, it is not normally necessary to change the start resistor for dual input voltage operation. Typically circuit has shown in figure

6  ANTISATURATION  TECHNIQUES  FOR HIGH-VOLTAGE  TRANSTSTORRS  INTRODECTION:  In high-voltage bipolar switching transistors, whereas the “fall time” (speed or dv/dt of the turn- off edge) is mainly determined by the shape of the base drive current (turn-off) characteristic. The storage time (delay between the application of the base turn-off drive and the start of the turn-off edge) is dependent on the minority carrier concentration in the base region immediately prior to turn- off action. To solve this problem we can use the “BAKER DIODE CLAMP” method.  Questions and Answers:  Q1:What would be the main advantage of using an ant saturation drive technique in high-voltage switching transistor?  This circuit has the advantage that, because it is an active drive clamp (with negative feedback). It compensates for the inevitable variations in gain and saturation voltage of the various devices. Also, it responds to change of parameters within the switching transistor that occur as a result of temperature and load variations.  Q2:Describe the action of a typical antisaturaion clamp circuit used for bipolar transistors.

7  Diodes D1 and D2, in series with the base drive to Q1, provide a voltage drop in addition to the transistor Vbe, so that the drive voltage at node A will rise to approximately 2V when Q1is driven on.  As Q1 turns on, the voltage on its collector will fall toward zero. When the voltage reaches approximately 1.3V, diode D3 will conduct and divert drive current away from the base and into the collector of Q1. As this clamping action is subject to negative feedback, it will self- adjust until the collector voltage is effectively clamped at 1.3V. As a result, the transistor is maintained in a quasi-saturated “on” state with just sufficient base drive current to maintain this condition. This on state maintains minimum minority in the base region during the “on” state, given minimum storage time during the turn-off current. Figure 1 shows a typical circuit. 

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9  START-UP METHODS  INTRODECTION:  If we only use the auxiliary supply to power the supply converter circuits, it will not be required when the converter is off. For the special case, the main converter transformer can have winding to provide the auxiliary power needs.  However, for this arrangement, some form of start-up circuit is required. Since this start circuit only needs to supply power for a short start-up period, very efficient start systems are possible. 

10  There are two methods for start up methods.  1: dissipative (passive) start circuit  2: transistor (Active) start circuit

11  TURN-ON VOLTAGE OVERSHOOT PREVENTION  INTRODUCTION:  When we switched on the power supply, either from the line input switch or by electronic means (say from a TTL logic “high” signal), there will be a delay while the power and control circuits established to their correct working condition.  TYPICAL CAUSES OF TURN-ON VOLTAGE OVERHOOT IN SWITCHMODE SUPPLIES:  In most switch mode power supplies, a controlled star-up sequence is initiated at switch- on. A turn-on is from a line input switch. The first action will be “inrush limiting,” where a resistive elements in series with the line input reduces the peak inrush currents for a few cycles while the input capacitors are charged-up.  After this, there will be a soft-start action. For this action the width of pulse power switching is progressively increased to establish the correct working conditions for inductors, transformers, and capacitors. The voltage on the output of capacitors established the require output voltage. However, it is still possible for the output overshoot. Figure shows the output filter and control amplifier of typical cycle- controlled switch mode power supply. The control amplifier has a simple pole-zero compensation networks to stabilize the loop.

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13  When the input voltage applied to this supply, and throughout the start-up phase, the control amplifier A1will recognizes the out put voltage as low. It gets the maxima out put pulse width from the ramp comparator A2. Now the A1will remain in saturated “high” state, with it is out put near +5V, and C1 will charge to +5V. During this period the pulse width will be under the control of A3. Therefore, the A3 will remain “high” until the out put voltage within 1 or 2mv. In this point the out put capacitor has been charged and considerable current will be established in the out put of inductor L1. After the voltage passes though a required value A1 will start to response. However, the delay will now ensue while the compensation network R1, C1 establishes its correct DC bias. Now the out put voltage of A1 is near +5V and the slew rate of the A1 is defined by the time constant of R1, C1. The correct amplifier working condition is not established yet. In this example the delay is approximately 500 us. During this period the out put of A1 must be close to 2.5V before to come under the control on pulse width of A2. This Daley, together with the excess current now flowing in the out put inductor L, will cause a considerable overshoot. Figure shows the out put voltage will go to 7.5V in this example.

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15  HOW TO PREVENT THIS OVERSHOOT:  By making the soft-start action very slow, allowing the amplifier to take over before the overshoot is too large we can reduce the overshoot. The linear power control circuit shown in figure In this circuit the 2.5V reference voltage for the control amplifier will be near zero at the no inverting input to the amplifier when first switched on, as C1 will be discharged prior to initial switch-on. The voltage on C1 will increase at a rate somewhat slower than the soft-start action. The control amplifier will establish its normal working condition at much lower out put. Figure is  Showing the typical circuit.

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18  Since the correct bais conditions for C2 and A1 were established at a much lower voltage, there will not be a overshoot. Figure is showing the turn on characteristics of modified circuit.