W-band InP/InGaAs/InP DHBT MMIC Power Amplifiers Yun Wei, Sangmin Lee, Sundararajan Krishnan, Mattias Dahlström, Miguel Urteaga, Mark Rodwell Department of Electrical and Computer Engineering, University of California tel: , fax

W-band MIMIC Power AmplifiersIMS2002 Y.C.Chen et. Al. IPRM, May stage 94 GHz W-band HEMT power amplifier 0.15  m composite-channel InP HEMT I max =750mA, V BR =7V, P out = 316 mW J. Guthrie et. Al, IPRM, May 2000 Cascode 78 GHz HBT power amplifier transferred substrate InGaAs/InAlAs SHBT I max =100mA, V BR =2.5V, P out = 12 mW This work single stage W-band HBT power amplifiers transferred substrate InP/InGaAs/InP DHBT I max =128mA, V BR =7V, P out = 40 mW Highest reported power for W-band HBT power amplifier

HBT processing Normal emitter and base processing  no collector contact polyimide isolation, SiN insulation, interconnection metals (M1 and M2), Benzocyclobutene planarization, thermal via and ground plane plating Flip chip bounding to carrier Substrate etching Schottky contact collector simultaneous scaling of emitter and collector widths  Wiring environment Micro strip transmission line  BCB dielectric,  r =2.7, t=5  m MIM capacitors  BCB bypass capacitor, SiN capacitor (  r =7, t=0.4  m ) NiCr resistor  R  =40  /  Low via inductance, reduced fringing fields, increased conductor losses Transferred-Substrate HBT MMIC technology

MBE DHBT layer structure Band profile at V be =0.7 V, V ce =1.5 V emitter InGaAs 1E19 Si 500 Å Grade 1E19 Si 200 Å InP 1E19 Si 900 Å InP 8E17 Si 300 Å Grade 8E17 Si 233 Å Grade 2E18 Be 67 Å InGaAs 4E19 Be 400 Å Grade 1E16 Si 480 Å InP 2E18 Si 20 Å InP 1E16 Si 2500 Å Multiple stop etch layers Buffer layer 2500 Å base collector substrate 400 Å InGaAs base 3000 Å InP collector

0.5  m Transferred-Substrate DHBT UCSB Sangmin Lee BV CEO = 8 V at J E = 0.4 mA/  m 2 f max = 462 GHz, f  = 139 GHz V ce(sat) ~1 V at 1.8 mA/  m 2

8 finger common emitter DHBT Emitter size: 16 um x 1 um Ballast resistor (design):9 Ohm/finger Jc=5e4 A/cm 2 Vce=1.5 V First Attempt at Multi-finger DHBTs: Poor Performance Due to Thermal Instability thermally driven current instability  collapse UCSB low f max due to premature Kirk effect (current hogging) excess base feed resistance ARO MURI

Multi-finger DHBTs: Design Challenges UCSB ARO MURI Thermal instability further increases current non-uniformity Ic Temperature collector SiN emitter contact basepoly BCB Metal strip Au Via Steady state current and temperature distribution when thermally stable Self-aligned base contact thickness=0.08  m base feed sheet resistance:  s =0.3  / significant for > 8 um emitter finger length Large Area HBTs: big C cb, small R bb,  even small excess R bb substantially reduces f max 0.08  m Emitter contact Metal1 Base contact Thermal instability (current hogging) in multi-finger DHBTs: Distributed base feed resistance: Ic Temperature

Large Current High Breakdown Voltage Broadband InP DHBT UCSB ARO MURI 8-finger device 8 x ( 1  m x 16  m emitter ) 8 x ( 2  m x 20  m collector ) 7  m emitter spacing ~8 Ohm ballast per emitter finger f max >330 GHz, V brceo >7 V, J max >1x10 5 A/cm 2 2nd-level base feed metal Ballast resistor emitter collector Flip chip

DHBT large signal model: Gummel-Poon model + BC parasitics + thermal feedback Gummel-Poon model: from DC characteristics Base-collector parasitics: from measured S-parameters Thermal effects: modeled by power-controlled base current & V be variation, using the measured thermal resistance  ta. Dynamic Load Lines: Ammeter is internal to Ccb; optimum HBT load has zero phase angle between ammeter and voltmeter UCSB instant -aneous power average power

Implementing the Thermal Feedback Large Signal Model in Agilent ADS UCSB

Large signal DHBT model -Comparison of DC and RF simulation and measurement measured simulation U h21 S21 S11S22 S12 UCSB

InP TS DHBT Power Amplifier Design ARO MURI Vce (V) Ic (A) I max V sat V CE_BR Designed using large signal model derived from DC-50 and GHz measurements of previous generation devices Output tuning network loads the HBT in the optimum admittance for saturated output power Shunt R-C network at output provides low frequency stabilization Electromagnetic simulator (Agilent’s Momentum) was used to characterize passive elements Low frequency stabilization Optimum admittance match Input match

W band 128  m 2 power amplifier UCSB common base PA 0.5mm x 0.4 mm, A E =128  m 2 ARO MURI f 0 =85 GHz, BW 3dB =28 GHz,G T =8.5 dB, P 1dB =14.5 dBm, P sat= 16dBm Bias: I c =78 mA, V ce =3.6 V

W band 64  m 2 power amplifier UCSB cascode PA 0.5mm x 0.4 mm, A E =64  m 2 ARO MURI f 0 =90 GHz, BW 3dB =20 GHz, G T =8.2 dB, P 1dB =9.5 dBm, P sat= 12.5 dBm Bias condition: I c =40 mA, V ce_CB =3.5 V, V ce_CE =1.5 V bias

Conclusions Wideband Power DHBT: I c = 100 mA, V ce =3.6 V, f max =330 GHz thermal design and base feed design critical for wide bandwidth Power DHBT large signal modeling Wideband Power amplifiers: f 0 =85 GHz, BW 3dB =28 GHz,G T =8.5 dB, P sat= 16dBm Future work Higher power DHBTs: lumped 4-finger topology and longer emitter finger Multi-stage wideband power amplifiers ~200 GHz power amplifiers Acknowledgement Work funded by ARO-MURI program under contract number PC UCSB IMS2002