1. DC/DC Converters for Automotive Applications; Systems Training
Colin Gillmor: (HPC)
Slide 1: This slide is for the 5 slide abstract ONLY
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2. Colin Gillmor Applications Engineer
Career
MEngSci, University College Cork, Ireland
PSU designer with Artesyn,
PSU Controller systems and Applications support with TI
Expertise
PSU System and Applications design
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My Bio
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3. Training summary TI Information - Selective Disclosure
DC/DC Converters for Automotive Applications; summary:
The demand for Electric Vehicles (EV) is increasing rapidly. This training session with a look at a typical EV system block diagram and then focus on the DC/DC applications within these systems.
Training level: Intermediate
Course Details:
Audience: All
Specific TI Designs & Parts Discussed:
TID #’s: PMP7246
Part #’s: UCC28951-Q1, LM4132-Q1 UCC Others
WEBENCH tools: N/A
What you’ll learn:.
Learn about a typical EV power system block diagram
Understand why they are designed in this way
How the Phase Shifted Full Bridge topology is used in EV applications
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Slide 2: This slide is for the 5 slide abstract ONLY
This slide is just a summary of the training – nothing in particular to note other than the text on the page
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4. Detailed agenda Electric Vehicle power systems – Block diagrams
Introduction to Battery Charging
Designing multi-kW power supply systems using the UCC28951-Q1
The Phase Shifted Full Bridge
High Power Battery Charger using the UCC28951-Q1
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Again – just read the slide
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5. Electric Vehicle power systems
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6. HEV/EV – Powertrain EE* overview
Engine Management
Gasoline & Diesel Engine ECU
Engine Actuators
Transmission
Manual, Automatic, & Shift-by-Wire Transmission
Transmission Actuators
HEV/EV
Battery Management
On-board Charger
Inverter
DC/DC Converter
Regenerative Braking
Powertrain Sensors
Pressure
Position
Temperature
Exhaust
Knock
Speed
Fluid Concentration/ Quality
Power Steering
Electric Power Steering
Hydraulic Power Steering
Steer-by-Wire
*EE = End Equipment
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Hybrid Electric Vehicle / Electric Vehicle.
This is an overview of some of the many applications for power electronics and sensors in electric vehicles.
This presentation will concentrate on some of the applications in the HEV/EV (red) section. In particular we’ll look at the on-board charger and DC/DC converter applications.
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7. HEV/EV DC/DC Converters
Bidirectional 48V-12V
Bidirectional HV-LV
Unidirectional 48V-12V
Unidirectional HV to LV – Analog Loop
On Board Charger
Unidirectional HV to LV – Digital Loop
Auxiliary Power Supplies
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This is a survey of the types of DC/DC converter used in HEV/EV End Equipments
There are different levels of electrification in use, ranging from micro Hybrid with simple automatic engine start/stop functions through to more complex Mild Hybrid, Full Hybrid to full EV (no Internal Combustion Engine) and a very wide range of power conversion requirements are found
Initial strategies simply move power hungry systems from 12V to 48V. For example an integrated starter/generator can be operated off 48V with reduced currents and less weight due to lighter cabling. Strategies like this allow what is called mild-hybrid operation, that is a combination of 12V and 48V electrics with a conventional Internal Combustion engine for traction. More complex, full hybrid solutions add regenerative braking systems such as used on the Toyota Prius and these require sophisticated bidirectional power inverters. Full Electric Vehicles move the entire power train to electric (Nissan Leaf and Tesla for example) and dispense completely with the Internal Combustion engine. In this case the traction system is run from high voltage (typically 400V) Li Ion batteries and lower voltages are used for other functions.
DC/DC converters are widely used for point of load and other applications at power levels from a few watts to multiple kW. Both isolated and non-isolated converters are used.
Bidirectional HV – LV converters are used to move energy between the onboard DC sources – 400V Li Ion, 48V and 12V Lead Acid batteries
Unidirectional HV –LV converters are used to charge 12V or 48V lead acid batteries, typically from a 400V source such as Li Ion batteries or the output of the on-board charger. Some applications require sophisticated controls for which digital control is appropriate and some may be controlled by an analog controller.
Hybrid and Full EV applications normally require bi-directional multi-phase inverters to drive the wheels and for re-generative braking.
On Board chargers convert line current (AC) to DC and provide an isolated DC output for battery charging. These modules are normally two stage converters, the input stage provides power factor correction with a 400V output. The second stage provides isolation and Constant Voltage / Constant Current operation needed for battery charging.
This presentation concentrates on high power isolated DC/DC converters – especially those used for battery charging purposes.

8. System Block Diagram External Link Vehicle Grid Connection EVSE* AC/DC
Battery
AC Power
Signalling
DC Power
This system is characterised by:
High Power levels
Dangerous voltages
Dangerous currents
Harsh environment
Proximity Sensor
GFCI*
SAE
Level 1: Single phase: AC power, 1.92kW
Level 2: Split phase: AC power, 19.2 kW
Level 3: DC power, 240kW
*EVSE – Electric Vehicle Service Equipment
*GFCI – Ground Fault Current Interruptor
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Here is a high level block diagram of an electric vehicle charging system. This takes energy from the AC grid and uses it to charge an on-board battery. We will consider two kinds of battery here, low voltage Lead Acid batteries operating at 12V or 48V and Lithium Ion batteries operating at around 400V.
There are various charger power levels – in the US these are described as Level 1, 2, 3 by the SAE (Society of Automotive Engineers. Their specifications refer of course to chargers connected to the US grid.
The grid source for Level 1 chargers will normally be a domestic socket – in the users home for example. The power drawn from the socket will be limited to about 1.92kW from a normal US grounded receptacle. This works well for overnight charging.
Level 2 refers to a connection to a single phase, dedicated 208/240Vac connection. Normally current limited to 32A. This means that Level 2 chargers operate at up to about 19.2kW. Charging time varies of course but will normally be in the range 1 to 4 hours.
Level 1 and Level 2 have an AC connection from the charging station to the vehicle being charged and the power level is relatively modest. Shorter charging times require higher power charging and to do this Level 3 uses a high voltage DC link to achieve charging times in the range 10 to 30 minutes. The PFC stage is now off board – ie not in the vehicle and the DC link is at up to 600V.
EVSE (Electric Vehicle Service Equipment) is a full implementation of a J1772-compliant service equipment specification.
The ‘link’ is just the cable from the charging station to the vehicle – fixed at the charging station end with a plug at the vehicle end. The plug is specially designed to be safe and in addition to the power transfer cables, contains a proximity sensor (switch) and signalling cable. The link ensures that power is not applied until the plug is correctly inserted in the socket. It also provides GFCI (Ground Fault Current Interruption) aka ELCB (Earth Leakage Current Breaker) and other safety functions. The proximity detector is used to prevent the car driving away while the cable is still connected. The signalling connection allows communication between the external EVSE and vehicle can communicate – for billing, power transfer level etc. Various ‘standard’ connectors exist – the US market uses the SAE - J1772 connector the European market uses the ITC Type2 (Mennekes) connector and there doesn’t appear to be much prospect of a common connector for AC but I think the connector standards aren’t fully settled yet and a standard DC (high power) connector may appear..
Inside the Vehicle is an AC/DC ‘on board’ charger. The higher power charging systems use a DC link and the AC/DC conversion is of course replaced by a DC/DC conversion
The battery being charged depends on the type of vehicle – in mild hybrid vehicles where there are no traction motors it may be a high capacity 12V or 48V lead acid battery or if electric traction motors are used then a 400V Li Ion battery is more usual. Energy storage varies from 3kWh to 135kWh depending on the vehicle and whether it is a hybrid or full electric vehicle.
The link to TIDUB87 will bring you to a Reference Design for Level 1 and Level 2 Electric Vehicle Service Equipment (EVSE)

9. On-Board Charger (OBC)
What is the On-board Charger?
What does this EE consist of?
An On Board Charger is used in an electric vehicle (EV) or hybrid electric vehicle (HEV) to charge the traction battery (48V or HV usually ~400V)
This includes:
Converts the grid 50/60Hz into DC
Adjusts the DC level to the levels required by the battery and provides the galvanic isolation
Usually includes a Power Factor correction (PFC)
PFC Controller and Rectification
High Efficiency rectification with lowest harmonic impact to the grid
Controller
Analog or Digital Control (<2kW to >100kW)
Adjusts the DC level to the levels required by the battery
Galvanic Isolation
Galvanic Isolation Grid to Battery
Bias Supply
Diagnostics
Temperature Sensing
Current & Voltage Sensing
Iso Barrier
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The On-Board charger takes energy from the grid and uses it to charge the on-board batteries. Ideally it will do this with high efficiency > 95% or so. This module will include Power Factor Correction to minimise harmonic currents drawn from the line. It will also include protection functions.
The input to the OBC is line current - single phase, 115V / 230V. Its purpose is to provide Power Factor correction and to convert the AC line voltage into a steady DC voltage – usually at around 400V – which is then fed into the input of a second stage DC/DC converter. For Safety purposes this DC/DC converter will include galvanic isolation to a high level (to IEC60950 for example).
This equipment is often – although not always – mounted in the engine compartment and it may be exposed to the very harsh environment that exists there. Liquid or air cooling is normally designed into the charger.
This equipment will also include built in diagnostics and communication (CAN)
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10. System Block Diagram
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This is an overview of an EV system – it’s a superset of all the potential power conversion opportunities in an EV and not all are present in any given EV.
There are a number of AC/DC opportunities for on board battery chargers – these are connected to wall plugs – in reality the higher power applications will be connected to a dedicated charging station via an EVSE as described before. Certainly some of the lower power applications may be connected to something like a domestic outlet – at the expense of an increased charging time of course.
Wireless charging has some attractive features – not least is the lack of a physical connection between the vehicle and the charger. There is also the possibility to add wireless charging points to normal parking spaces so that they become part of the overall infrastructure – in supermarket or workplace car parks for instance. This has the added attraction that the vehicle is charging while the user is doing something else, not simply waiting around while the vehicle is charged. But this is outside the scope of this presentation and I’m not going to say anything more about wireless charging here.
There is a large number of motor drive opportunities. Inverters to run BLDC motors in the Air Conditioning, Seat positioning, Water pump, Power Steering and Cooling systems for example. The highest power levels are seen in the traction motors used propulsion and for re-generative braking. High power bi-directional DC/DC and AC/DC converters are used in this application. Again these are outside the scope of this presentation.
A large number of isolated and non-isolated dc/dc converters are used for point of load and other control electronics – these are typically low power applications.
This presentation is going to concentrate on some of the unidirectional DC/DC applications at medium to high power 1kW to 3kW and to show which topologies are chosen and how they are designed for battery charging purposes.

11. System Block Diagram Potential DC/DC Applications for UCC28951-Q1
(green)
Unidirectional
High Power
ZVS for low loss on HV inputs
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The applications highlighted in green are ones suited to a Phase Shifted Full Bridge operating under analog control. The UCC28951-Q1 is the controller of choice for these applications.
As we will discover later – the PSFB is well suited to high voltage, high power dc/dc applications where the output voltage can vary over a wide range. In this case, the output is fed into a battery whose voltage can vary over a wide range – up to 2:1 in some cases.
The PSFB topology uses four switches on the primary and the associated complexity is normally only justifiable in high power applications where the improved core utilisation of the full bridge is needed for efficient operation.
The other thing to note is that all of these applications take a high voltage DC input – typically around 400Vdc and at these voltages, the ZVS action of the PSFB significantly reduces switching losses in the primary side MOSFETs.
As we will see later, the UCC28951-Q1 is well suited to the control of these unidirectional DC/DC converters

12. Introduction to Battery Charging
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13. Lead Acid battery TI Information - Selective Disclosure
Chargers operate in CI and CV modes
Compensation for battery temperature
12V / 48V nominal battery voltages are common
2.35V per cell (typ) when charged
1.9V per cell (typ) when discharged
33-42 Wh/kg energy density
Battery damage if not fully charged – Sulfation
Float charge compensates for self-discharge
Ideal float voltage is a function of temperature
Deep discharge damages battery
≈70%
98%
100%
Lead Acid
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Lead Acid batteries have been in existence for over 150 years and have been the main type of rechargeable battery in use for most of that time. Despite its age the technology remains popular and it has been much improved since it was originally invented in It’s still in widespread use in cars for example. 12V and 48V types are common in EV applications and they offer a reasonable energy density in the range 33 to 42Wh/kg. And a good cost performance ratio.
Chargers for Lead Acid batteries must operate in Constant Current and Constant Voltage modes. Initial charging is at a constant current and the battery terminal voltage increases steadily to almost full voltage. At this point the battery is about 70% charged. This part of the cycle takes about 40% of the total cycle time. The charger then changes to constant voltage to supply the remaining 30% or so of charge. Once fully charged (98%) the voltage is dropped to the so called float level. This compensates for the batteries’ tendency to self discharge can be maintained indefinitely.
The constant voltage applied during the Absorption charge phase is a function of temperature and the precise battery chemistry involved. This means that the battery temperature must be monitored and the charger voltage adjusted accordingly.
Lead Acid batteries should be kept fully charged when not in use.
Lead acid batteries cannot be ‘fast charged’ and a full charge takes 12 to 16 hours.
The negative terminal can sulphate causing permanent damage if they are not fully charged
They can outgass and the positive plate can corrode if they are overcharged
They can also be permanently if fully discharged – so called deep discharge.
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14. Lithium Ion battery TI Information - Selective Disclosure
400V nominal battery voltage
300V minimum when discharged
Wh/kg energy density
Very tight end voltage tolerances
Typ 4.2V/cell ±50mV (±1.2%) → OVP
Over Charging can damage battery
Temperature rise during charging → OTP
Battery pack cell balancing – not considered here
85%
100%
Lithium Ion 3%
Tradeoff
Charge Rate vs Charge Time vs Battery Life
Final Charge vs Battery Life vs Range - Fully charging Li Ion battery can reduce lifetime
70% charge / 20% discharge cycle for extended lifetime – but reduced range.
Preconditioning phase if deeply discharged – not shown here
Periodic ‘top up’ charge – not a continuous trickle charge
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Lithium Ion batteries were first proposed in the 1970s and developed into practical form in the 1980s. Today they are in widespread use in consumer electronics – laptops, mobile phones etc. Their main use in electric vehicles is as an energy source for the traction motors. In traction applications several thousand individual 4.2V cells are arranged in series parallel circuits and the resulting battery pack has a typical nominal output voltage of around 400V and a capacity which varies from 18 kW hr to 100kW hr. Management of the state of charge of individual cells in the battery to equalise charging among them and to work around ‘weak’ cells is complex and outside the scope of this presentation.
Li Ion batteries are much more sensitive to charging voltage than lead acid types and battery chargers must be able to hold to very tight voltage tolerances during the ‘Saturation’ and ‘Top up’ charge phases. There is no ‘float’ charging phase associated with this chemistry, instead a periodic top up charge is applied when the terminal voltage drops. Over charging can damage the battery.
These batteries may be charged more quickly than Lead Acid types – typically in 2 to 3 hours
There is a complex trade-off to be made between the charge rate, the charge time and battery life. In fact a fast charge can charge the battery to about 70% in a short time although the time to 100% charge is extended so that total charge time remains more or less constant. In fact, charging to 100% can overstress the battery and reduce its lifetime so the optimum charge level seems to be between 70% and 85%. The price for this if course is less available energy in the battery and reduced driving range.
Deeply discharged Li Ion cells must be preconditioned with a low constant current until their terminal voltage rises to a point where the full fast charge current may be applied.
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15. Battery Charger output regulation
Two regions
Constant Voltage regulation
Regulation down to zero current
Tight regulation tolerances
Temperature dependence
Constant Current regulation
Regulation down to approx half nominal Vout
If Vo drops below this –
Micro controller decides action
Li Ion battery ‘top up’ behaviour and final charge levels determined by MCU – this is a system level decision and trades stored charge against lifetime.
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Chargers for Lead Acid and Li Ion batteries are very different in their detail design but they do have some things in common and this slide shows some of these common features.
Basically these chargers must operate in both Constant Current and Constant Voltage modes depending on the state of charge of the battery. Initial charging is done in constant current mode – of course the current must be set to match the battery capacity and chemistry but that is outside the scope of this discussion. Normally we would expect CI operation in the range from max Vout at which point the charger switches to CV mode down to about 50% of max Vout. If the battery voltage is less than that, then depending on the chemistry it may be permanently damaged or need a special low current initial charging current. In any case we won’t consider that operating region any further because – in order to extend their operating lifetime (number of charge/discharge cycles) traction batteries in particular are not normally discharged fully.
Constant Voltage operation must be provided at all current levels from the maximum where it enters CC operation down to zero current when it is operating in float mode (Lead Acid) or in top up mode (Li Ion)
The charger must switch automatically between CI and CV modes. In CV mode, tight regulation with temperature compensation of the output voltage is normally necessary.
The UICC28951-Q1 does not provide temperature compensation so this feature must be supplied by an external temperature sensing network. We assume that this is done by an external micro controller which can sense the temperature and then adjust the CV operating reference according to the type of battery being charged.
In Li Ion applications, the microcontroller will also decide the ‘top up’ behaviour, turning the charger on and off as necessary. This is a system level behaviour and is chosen to satisfy a tradeoff between stored charge and battery lifetime.
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16. Designing multi-kW power supply systems using the UCC28951-Q1
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17. Systems Overview
Problem: Electric vehicles need systems to convert AC power into DC for storage in high (HV) and low voltage (LV) batteries and to convert the stored energy back to AC to drive the Motors. We’ve seen the overall system block diagram now we will examine how to design the DC/DC link between the PFC stage output and HV Li Ion and LV Lead Acid batteries.
Solution: The UCC28951-Q1 is a sophisticated device that controls the PSFB stage to achieve high efficiency at high power levels in conjunction with other TI devices.
Description of system solution: The PSFB power system is a key component of this system, specifications are shown in the next slide.
Key components: Texas Instruments offers a wide variety of devices for PSFB applications in H/EV. A few examples: The UCC28951-Q1 PSFB controller. The multi channel UCC kV isolated gate driver. The LM4132-Q1 Reference with 0.05% accuracy. INA520-Q1 and INA199-Q1 Current sense amplifiers, TLV316-Q1 op-amp
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Electric vehicles need systems to convert AC power into DC for storage in high (HV) and low voltage (LV) battery systems and to convert the stored energy back to AC to drive the Motors. Now we will examine how the DC/DC link between the PFC stage output and HV Li Ion and LV Lead Acid batteries is designed
The design uses a PSFB stage controlled by the UCC28951-Q1 to achieve high efficiency at high power levels. Other devices are used to perform ancilliary functions – references, current sensing and op-amps for example.
The PSFB power system is the key component of this system and a typical specification will be shown in the next slide.
Texas Instruments offers a wide variety of devices for use in Hybrid / EV applications. A few examples: The UCC28951 PSFB controller. The UCC kV multi channel isolated gate driver. The LM4132 Reference with 0.05% accuracy. INA520 and INA199 Current sense amplifiers, TLV316 op-amp etc. etc.

18. Example applications Phase Shifted Full Bridge UCC28951-Q1
12V Lead Acid battery charger
Input from PFC stage, Output charges battery
Min
Nom
Max
Vin
370V
390V
410V
Hold Up time
n/a
Vout 8V
12V
15V
Power Out
1kW
Modes
CI, CV, Float
Battery
Lead Acid
Max Iout
83A
Temp comp
Ext
400V Li-Ion battery charger
Input from PFC stage, Output charges battery
Min
Nom
Max
Vin
370V
390V
410V
Hold Up time
n/a
Vout
300V
400V
420V
Power Out
3.3kW
Modes
CI/CV/OFF
Battery
Li Ion
Max Iout
8.25A
Temp comp
Ext
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Here are typical specifications for two battery charger applications. A 12V Lead Acid battery charger at 1kW and a 400V Li Ion battery charger at 3.3kW. Their input is a 390V rail generated at the output of a preceeding PFC stage. These are typical specifications and of course real world specifications may differ in detail but the overall design approach will remain more or less unchanged.

19. The PSFB in multi-kW power supply systems
The PSFB is the topology of choice for high input voltage, high power applications because:
It achieves Zero Voltage Switching (ZVS) which significantly reduces switching losses.
It uses the full flux swing available from the transformer core so that a smaller transformer is possible.
The transformer primary is driven with the full input voltage minimising primary currents.
Efficiencies of greater than 99% can be achieved.
The main disadvantage is that it requires four active switches on the transformer primary.
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One of the first decisions in any design is to choose the topology – in this case we will use the Phase Shifted Full Bridge.
This topology is a good choice at these power levels and voltage because
It achieves Zero Voltage Switching (ZVS) which significantly reduces switching losses. This is especially important at high input voltages.
It uses the full flux swing available from the transformer core so that a smaller transformer is possible.
The transformer primary is driven with the full input voltage minimising primary currents.
Efficiencies of greater than 99% can be achieved – with careful design !.
The main disadvantage is that it requires four active switches on the transformer primary however the added complexity is justifiable at high power levels because of the improvement in transformer utilisation.

20. Phase Shifted Full Bridge
Active Leg
QA, QB
Passive Leg
QC, QD
Four switches, transformer, two rectifiers, inductor
Double ended topology
Buck like output stage
Four switching states per cycle
Two power transfer
Two freewheeling
Four ZVS transitions per cycle
Phase between legs controls conversion ratio
Complex control, requires IC
High power (1kW and upwards)
Can achieve zero voltage switching
Important for high Vin applications
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This is the schematic of a PSFB circuit
QA, QB, QC and QD are the four primary side switches. QE and QF are the secondary side rectifiers. Other secondary rectification circuits are possible, and we will look at these later but for now we will concentrate on this centre tapped full wave rectifier.
It’s a double ended topology which means that both ends of the transformer primary are alternately switched between Vin and 0V, this allows symmetrical positive and negative flux swings in the transformer core so core reset is not an issue. The transformer primary sees the full input voltage so currents are less than in other topologies.
The PSFB is a buck derived topology so the input / output conversion is directly proportional to the duty cycle.
There are four switching states and four ZVS transitions per cycle and we will look at these in more detail shortly.
There are different terms used by different sources when describing this topology. I’m going to use the convention usually used within TI – which is the one shown here.
Just to restate – the PSFB is well suited to high input voltage, high power applications because of excellent transformer core utilisation and its ability to achieve ZVS.
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21. Phase Shifted Full Bridge
Active Leg
QA, QB
Passive Leg
QC, QD
Can achieve zero voltage switching
ZVS and reduced cross conduction requires:
Dead time between QA OFF and QB ON
Dead time between QC OFF and QD ON
Reduced body diode conduction requires
Dead time between QA OFF and QF OFF
Dead time between QB OFF and QE OFF
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The PSFB can achieve ZVS BUT it is not GUARANTEED to do so. The ZVS transition is a resonant transition whose frequency is set by the capacitances and inductances at the switching nodes A and B above.
The system must be designed with enough energy storage to force the ZVS transition and the appropriate dead times between switching events to allow the transition to happen.
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22. Phase Shifted Full Bridge
Active Leg
QA, QB
Passive Leg
QC, QD
𝑉 𝑂𝑈𝑇 =𝐷 𝑉 𝐼𝑁 𝑁 𝑆 𝑁 𝑃
Buck Derived topology
OUTA, OUTB – reference pair
D controlled by phase shifting OUTC & OUTD
QE, QF are SRs, Diode rectification is possible
Mouse over the waveforms to play the animation
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The graphic shows the gate drives to QA and QB (red and blue) on top. QC and QD (red and blue) in the middle.
The voltage across the transformer is shown in red at the bottom.
These waveforms are all at 50% duty cycle – less 1% or 2% to allow time for ZVS transitions.
OUTA and OUTB are the reference pair, they don’t change.
OUTC and OUTD are phase shifted with respect to the reference pair to control the duty cycle.
Power transfer from primary to secondary happens when ‘diagonal pairs’ of switches are on. – QA and QD or QB and QC
The transformer duty cycle is simply the proportion of the switching cycle for which voltage is applied to the transformer
I’m going to play the animation now –
Mouse over the waveform graphic, press the ‘start’ button and let the animation speak for itself
You can press the pause button if you wish or replay the animation.
You can see how the duty cycle changes as the phase between QA/QB and QC/QD alters

23. Timing Diagram: Energy Transfer
QA, QD, QF are ON: others are OFF
First energy transfer interval
I_PRI is Iout /N* + Imag.
QF current is Iout
Current flow in red (pri) and blue (sec) paths
Currents at end of interval, solid red / blue
*N is the turns ratio
I_LOUT: increasing
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The next 8 slides will describe a typical switching cycle of the PSFB
Here we see the initial condition
The red and blue arrows indicate current flow in the primary and secondary circuits
QA and QD are on for a time Ton, which of course is determined by the control loop.
This puts a positive voltage across the transformer primary and the primary current is increasing.
The voltage at the secondary of the transformer is rectified by the SR QF.
And the current in the output inductor is increasing as energy is transferred from primary to secondary.

24. Timing Diagram: ZVS QA, QD, QF are ON: QC is OFF QD turns OFF
Node B charges to Vin as I_PRI current moves out of QD and into QC Body Diode*
QC: turns ON
QD turns off
DELCD – allows time for Node B transition
QC turns on at 0V (ZVS)
Leakage Inductance
L_lk
Current in QD goes to zero during interval
Uses L_lk energy.
Faster than Node A transition, because I_LOUT is at maximum
*ZVS transition
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At some time the PWM controller terminates the on time by turning QD off
DELCD is the the dead time between QD turning off and QC turning on. This dead time allows time for the energy stored in the leakage inductance to charge the capacitance at node B and this node swings positive from 0V when QD is turned off. The resonant transition is clamped when the body diode of QC turns on.
QC channel is turned on at the end of the DELCD interval but it does so with zero volts across it and no switching loss.
The energy available to drive this transition is greater than that driving the transition at node A later in the cycle. This means that this transition happens in a shorter time than the one at node A. This is the reason why the UCC28951-Q1 allows the designer to set different DELAB and DELCD times.
The primary and secondary currents at the end of the ZVS transition interval are shown in red and blue

25. Timing Diagram: Freewheeling
QA, QC, QE, QF are ON: others are OFF
T1 Primary is short circuited, VXFMR = 0V
T1 Sec is short circuited by QE & QF
Output current supplied by Lout
Current flows asymmetrically in T1 Sec !
QE turns on
½ ΔI_Lout + Iout
½ ΔI_Lout
QE turns ON
Secondary is shorted
I_LOUT: decreasing
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At the end of the resonant transition, QE is turned on. At this time both the primary and secondary of the transformer are short circuited.
So there is no voltage across the transformer primary and we know that V = L di/dt so if V equals 0 then di/dt must also equal 0. This means that the circulating current in the primary is preserved and is available to drive a ZVS resonant transition at the end of this interval.
This is the key feature of the PSFB that differentiates it from the PWM full bridge.
Output current is supplied by Lout through the transformer secondary. This current flows asymmetrically in the transformer because the secondary short circuit means there is no voltage available to force current out of one winding into the other. The average current remains in the QF and only the change in inductor current splits equally. This asymmetry means that RMS losses in the transformer secondary are higher than they would be if the currents shared equally.
This interval ends when the PWM clock starts the second energy transfer cycle -

26. Timing Diagram: ZVS QA, QC, QE are ON: QB is OFF QA turns OFF
DELAB – allows time for node A transition
QA, QC, QE are ON: QB is OFF
QA turns OFF
Node A charges to GND as I_PRI current moves out of QA and into QB Body Diode
QB: turns ON
QA turns off
QF turns off
after DELEF
DELEF
Leakage Inductance
QB turns on at 0V (ZVS)
Uses L_lk energy.
Slower than Node B transition, because I_LOUT is at minimum
QF turns OFF
Removes sec short
TI Information – Selective Disclosure 26
The PWM clock starts the second energy transfer cycle by initiating a second ZVS transition when it turns QA off
As with the earlier ZVS transition, the dead time between QA turning off and QB turning on, DELAB in this case, allows time for the energy stored in the leakage inductance to charge the capacitance at node A and this node swings negative from Vin when QA is turned off. The resonant transition is clamped when the body diode of QB turns on.
QB channel is turned on at the end of the DELAB interval but it does so with zero volts across it and no switching loss.
The energy available to drive this transition is less than that driving the previous transition at node B. This means that this transition takes longer than the one at node B. This is the reason why the UCC28951-Q1 allows the designer to set different DELAB and DELCD times.
QF is turned off during this interval and this releases the short circuit on the transformer secondary in readiness for the second energy transfer interval -

27. Timing Diagram: Energy Transfer
QB, QC, QE are ON: others are OFF
Second energy transfer interval
I_PRI is Iout /N* + Imag
QE current is Iout
Current flow in red (pri) and blue (sec) paths
QC turns off
*N is the turns ratio
I_LOUT: increasing
TI Information – Selective Disclosure 27
This is the second energy transfer interval during the switching cycle.
In this slide, QB and QC are on for a time Ton which is determined by the control loop.
The transformer polarity is now opposite to that in the first energy transfer interval and there is a negative voltage across the transformer primary and the primary current is increasing in amplitude.
The voltage at the secondary of the transformer is rectified by the SR QE and the current in the output inductor is increasing as energy is transferred from primary to secondary.
You can also see that the switching frequency seen by the output inductor is twice seen by the transformer –
this can cause confusion unless you are very clear about what you mean when you describe the switching frequency of a PSFB.

28. Timing Diagram: ZVS QB, QC, QE are ON: QD is OFF QC turns OFF
Node B charges to Gnd as I_PRI current moves out of QC into QD Body Diode*
QD: turns ON
QC turns off
DELCD –allows time for node B transition
Leakage Inductance
*ZVS transition
QD turns on at 0V (ZVS)
TI Information – Selective Disclosure 28
The PWM controller terminates the on time by turning QC off
As before the dead time allows time for the energy stored in the leakage inductance to charge the capacitance at node B and this node swings negative from Vin when QC is turned off. The resonant transition is clamped when the body diode of QD turns on.
QD channel is turned on at the end of the DELCD interval but it does so with zero volts across it and no switching loss.

29. Timing Diagram: Freewheeling
QB, QD, QE, QF are ON: others are OFF
T1 Primary is short circuited, VXFMR = 0V
T1 Sec is short circuited by QE & QF
Output current supplied by Lout
Current flows asymmetrically in T1 Sec
QF turns on
½ ΔI_Lout + Iout
½ ΔI_Lout
QF turns ON
Secondary is shorted
TI Information – Selective Disclosure 29
At the end of the resonant transition, QF is turned on. At this time both the primary and secondary of the transformer are short circuited.
As before (when QA and QC were both ON) there is no voltage across the transformer primary and the circulating current in the primary is preserved and is available to drive a ZVS resonant transition at the end of this interval.
As in the earlier freewheeling interval, current is supplied by Lout through the transformer secondary although, as before the current flows asymmetrically in the transformer
This interval ends when the PWM clock starts the second energy transfer cycle –
Output current is supplied by Lout through the transformer secondary. This current flows asymmetrically in the transformer because the secondary short circuit means there is no voltage available to force current out of one winding into the other. The average current remains in the QF and only the change in inductor current splits equally. This asymmetry means that RMS losses in the transformer secondary are higher than they would be if the currents shared equally.

30. Timing Diagram: ZVS QB, QD, QF are ON: QA is OFF QB turns OFF
DELAB - allows time for Node A transition
QB, QD, QF are ON: QA is OFF
QB turns OFF
Node A charges to Vin as I_PRI current moves out of QB into QA Body Diode
QA: is turned ON
QB turns off
QE turns off
after DELEF
DELEF
Leakage Inductance
QA turns on at 0V (ZVS)
Uses L_lk energy.
Slower than Node B transition, because I_LOUT is at minimum
QE turns OFF
Removes sec short
TI Information – Selective Disclosure 30
The PWM clock starts the next energy transfer cycle by initiating a ZVS transition by turning QB off
As before the dead time, DELAB, allows time for the energy stored in the leakage inductance to charge the capacitance at node A and this node swings positive from 0V when QB is turned off. The resonant transition is clamped when the body diode of QA turns on.
QA channel is turned on at the end of the DELAB interval but it does so with zero volts across it and no switching loss.
QE is turned off during this interval and this releases the short circuit on the transformer secondary.
This completes the switching cycle and the process repeats indefinitely.

31. Phase Shifted Full Bridge – reminder !
Active Leg
QA, QB
Passive Leg
QC, QD
𝑉 𝑂𝑈𝑇 =𝐷 𝑉 𝐼𝑁 𝑁 𝑆 𝑁 𝑃
Buck Derived topology
OUTA, OUTB – reference pair
D controlled by phase shifting OUTC & OUTD
QE, QF are SRs, Diode rectification is possible
TI Information – Selective Disclosure 31
Just a reminder of how the phase relationship between the OUTA/OUTB and OUTC/OUTD pairs controls the duty cycle.
Remember, OUTA/OUTB are the reference pair, OUTC/OUTD move around to control the phase.

32. On Board Charger < 3.3kW (UCC28951 Control)
MCU for system supervision
TI Information – Selective Disclosure 32
Here’s block diagram showing how a PSFB battery charger can be designed
The power train is controlled by a UCC28951-Q1 controller.
A microcontroller (MCU) provides system supervision functions –
MOSFET and Battery temperature monitoring
Output voltage temperature compensation
Float level setting (Lead Acid)
Top Up charge control (Li Ion) etc etc.
An isolation amplifier monitors the input voltage and its output may be used by the micro controller (MCU) to provide an input UVLO
The current transformer provides the current sense signal (CS) needed for Peak Current Mode control
QA, QB, QC, QD are the four primary side switches and QE and QF are the SRs on the secondary.
Isolated Drivers are used to drive the primary side devices. Low side MOSFET drivers are used to drive the secondary side SRs.
Precision references are needed to achieve the tight output voltage tolerances needed for battery charging – especially for Li Ion chargers.
An output current sensing network and error amplifier is needed to provide accurate CI regulation
Output voltage regulation is achieved in the usual way with an external error amplifier and precision reference.
The MCU controls the output voltage by using a digital potentiometer
Various auxiliary power supply rails are needed and these are supplied by some low power flyback PSUs.
TI Information - Selective Disclosure

33. On Board Charger: Sec Bias Flyback
Small Flyback PSU for Secondary side power
UCC28700-Q1 for example
Primary side regulation – no need for an optocoupler
Simple, low cost transformer
Small size, 6 pin SOT23
Efficiency probably about 75%
power level is low – estimate 5W
Variable frequency – as with all DCM flyback devices
Cable compensation (CBC) probably not needed – tie CBC pin to GND
Design tools available
Webench
Reference designs
Evaluation Modules
12V output
TI Information – Selective Disclosure 33
The secondary bias rails are provided by a small flyback psu.
Here we show a design using the UCC28700-Q1 primary side regulator
This would normally provide a 12V output at low current < 500mA or so
This device allows a low cost design, a simple low cost transformer and a small overall design.
TI provides design tools, WebBench support, reference designs and evaluation modules to use with this part.
TI Information - Selective Disclosure

34. On Board Charger: Pri Bias Flyback
Small Flyback PSU for Primary side power
UCC28700-Q1 for example
Primary side regulation – no need for an optocoupler
Simple, low cost transformer
Small size, 6 pin SOT23
Efficiency probably about 75%
power level is low – estimate 5W
Variable frequency – as with all DCM flyback devices
Cable compensation (CBC) probably not needed – tie CBC pin to GND
Design tools available
Webench
Reference designs
Evaluation Modules
12V output
TI Information – Selective Disclosure 34
This is the same device as used for the Secondary Bias Flyback, the main difference is that only functional insulation is needed in the transformer rather than the safety critical reinforced insulation needed for the secondary bias.
TI Information - Selective Disclosure

35. On Board Charger: Isolated Driver, Option 1
Primary/Secondary Isolation
Switching of Primary side MOSFETs
High Side and Low Side outputs needed
4 isolated outputs in total
2 high side drives, 2 low side drives
Isolation to 5.7kVRMS
2 x UCC21520-Q1, 4A, 6 A driver
Low Propagation Delays
Good Propagation Delay Matching
Adjustable Dead Time
Safety Features, UVLO etc.
As with all drivers, PCB layout is critical
TI Information – Selective Disclosure 35
The drive signals for the primary side MOSFETs originate in the controller located on the secondary side.
The UCC21520-Q1 isolated driver can be used to cross the isolation barrier and to drive the MOSFETs directly.
Each driver has two independent outputs which may be used to drive both low side (QB and QD) and high side (QA and QC) switches.
Propagation delays are short and well matched – in this application the adjustable dead time feature is not needed and may be disabled (dead times are controlled by the UCC28951-Q1).
The PCB layout around the drivers is critical and is described in detail in the device datasheet.
TI Information - Selective Disclosure

36. On Board Charger: Isolated Driver - Option 2
Pri/Sec Isolation
ISO7740-Q1 provides pri/sec isolation
5kV RMS
0/10V signal from UCC28951-Q1 needs attenuation (2:1) to meet ISO7740-Q1 input level.
Gate drivers drive the MOSFETs
ISO7740FQDWQ
F option – outputs default LOW !
High
Side
Low
Side
ISO7740F-Q1
TI Information – Selective Disclosure 36
An alternative method of driving the primary side MOSFETs is shown here.
The ISO7740F-Q1 is a digital isolator device. It takes a 5V signal at its inputs, so some attenuation of the 10V OUTx signals from the UCC28951-Q1 is needed a resistive divider for example.
If this device is used, it’s important to use the F option where the outputs default low in the event of a loss of input signal.
Separate MOSFET drivers are needed to interface the digital signals from the isolator with the MOSFETs. This is not always a disadvantage. Separating the isolation and driver functions means that the drivers can be placed close to the MOSFETs without having the constraints due to the primary / secondary isolation barrier and the associated PCB creepage and clearance distances.
TI Information - Selective Disclosure

37. On Board Charger: Rectification – General
Choice of secondary rectification depends on -
Output Voltage
Output Current
400Vout:
Diodes – Simple solution, a good choice for 400V
Full Wave or Bridge options
Reverse recovery losses makes SiC a good choice
12Vout:
SR – Good option at 12V out, body diode reverse recovery losses can be significant
Full wave with centre tap or Bridge with single secondary winding options
SRs require a MOSFET driver
Schottky diodes might be an option higher losses but easier drive and no reverse recovery problems
Current doubler with SR is a good option – single sec. winding
TI Information – Selective Disclosure 37
Up till now we have assumed that a full wave rectifier with a centre tapped secondary and synchronous rectifiers is used – like the circuit shown here –
In fact there are a few alternative rectifier circuits that can be used and the choice depends on the output voltage and current levels
At 400Vout:
Diodes, especially Silicon Carbide ones are a simple solution
Full Wave or Bridge options – tradeoff of rectifier losses and transformer secondary complexity
At 12Vout:
Synchronous Rectifiers are a sgod option at 12V out, body diode reverse recovery losses can be significant
Options to go for Full wave with centre tap or full Bridge with a single secondary winding
SRs require a MOSFET driver of course such as the UCC27424-Q1 shown here.
Schottky diodes may be an option. The losses due to the forward voltage drop are higher but easier drive and no reverse recovery problems
A Current doubler secondary with SR is a good option – it uses a single winding on the transformer and splits the current between two inductors
Now we will have a brief look at some of these alternatives
TI Information - Selective Disclosure

38. On Board Charger: Rectification – 12V output
SRs are large rectifier MOSFETs.
UCC27424-Q1 is a dual non-inverting MOSFET driver.
MOSFETs see 2 x Vin_max Ns/Np + margin
Use 30V devices for 12V output
Reverse recovery losses in SR can be significant
Centre tapped secondary
Half of sec winding ‘idle’ at a given time
‘Idle’ half may cause proximity losses
TI Information – Selective Disclosure 38
This is the full wave rectifier with a centre tapped secondary.
MOSFETs are used as synchronous rectifiers
The UCC27424-Q1 is a dual non-inverting MOSFET driver with 4A drive capability.
The MOSFETs see a voltage stress of 2 x Vin_max Ns/Np and we must increase this to allow for switching spikes and derating margins
We should be able to use 30V devices for a 12V output
Reverse recovery losses in SR can be significant. These losses must be minimised by careful selection of the SR device and switching timing.
The Centre tapped secondary means that half of sec winding is ‘idle’ at a given time and the transformer winding structures must be designed carefully to minimise proximity losses in the secondary – including losses in the ‘idle’ half.
TI Information - Selective Disclosure

39. On Board Charger: Rectification – 12V output
Current Doubler output with Schottky Rectifiers
Current Doubler – suited to high current outputs
Requires Current Mode Control
Ripple current cancellation in Cout
Single winding on transformer secondary
best use of transformer winding window
Two output inductors needed
Each inductor carries half the output current
Vf losses are significant – depends on diode
Heatsinking requirements significant
Electrically – this is the simplest option
Significant losses in Diodes.
Secondary
Centre Tapped
Current Doubler
Ind Current
I_out
I_out/2
Ind Freq
2 fSW
fSW
Inductance
L_out
<Lout*
* Depends on Duty Cycle
TI Information – Selective Disclosure 39
This is the Current Doubler output circuit shown here with Schottky Rectifiers.
This topology is well suited to high current outputs
Requires Current Mode Control to force equal currents in the two output inductors
An advantage of this topology is that there is a significant amount of ripple current cancellation in Cout
The single winding on transformer secondary simplifies its design and makes best use of the transformer winding window
A disadvantage is that two output inductors needed but each inductor carries half the output current which means it can be made lighter
Vf losses are significant – depends on diode
Electrically – this is the simplest option
TI Information - Selective Disclosure

40. On Board Charger: Rectification – 12V output
Current Doubler output with Synchronous Rectifiers
MOSFETs see 2 x Vin_max Ns/Np + margin
Reverse recovery losses in SR can be significant
SRs are ground referenced – simple driver
UCC28951-Q1 OUTE and OUTF signals are driver inputs
May need to parallel several MOSFETs
Use separate gate drives
or separate gate drive resistors
Needs careful layout to avoid HF oscillations
TI Information – Selective Disclosure 40
This is a Current Doubler output with Synchronous Rectifiers
It has much in common with the earlier diode rectified Current doubler circuit
As before the MOSFETs see 2 x Vin_max Ns/Np and design margin will have to be added to this
Reverse recovery losses in SR can be significant – the MOSFETs need to be chosen carefully
The SRs are ground referenced which allows the use of a simple driver like the UCC27424-Q1
Depending on the current levels the designer may need to parallel several MOSFETs
If you do this, then use separate gate drivers or separate gate drive resistors
Needs careful layout to avoid destructive HF oscillations – but it can be done !
TI Information - Selective Disclosure

41. On Board Charger: Rectification – 12V output
Full wave rectification with SR
Simplest transformer
Single secondary winding
Single output inductor
Two SR voltage drops in current path
SRs see Vin_max Ns/Np + margin
Reverse recovery effects in SR diodes
SR drive complexity
2 low side drives, 2 high side drives
TI Information – Selective Disclosure 41
This is a full wave rectified secondary.
The transformer secondary has a single winding and uses a single output inductor
The transformer secondary current is highest in this configuration but the voltage stresses in the SRs are half those in the Current Doubler circuti.
The main disadvantage of this configuration is that four SRs are required. In addition two high side and two low side SR drivers are needed. In fact this circuit is rarely used,
except of course in bidirectional converters which are outside the scope of this presentation.
TI Information - Selective Disclosure

42. On Board Charger: Rectification – 400V output
SiC diodes are simplest solution
Positive temp coefficient of Vf
Relatively low currents allow use of centre tapped secondary
V stresses on diodes are 2 x Vin_max Ns/Np + margin
Use 1200V rated SiC diodes
Full Bridge rectification
Halves V stresses
Simplifies secondary
Increases rectifier losses
Infineon IDH10G120C5
TI Information – Selective Disclosure 42
Silicon Carbide Diodes are the simplest solution for on a 400V output. Either a full wave rectifier shown at the top right or a full bridge rectifier can be used with the normal tradeoffs of reverse voltage stresses, transformer complexity and losses due to the forward voltage drops in the diodes.
SiC diodes have a complex Vf versus temperature characteristic. It is positive when the current is highest but negative at lower currents. This makes current sharing by paralleling diodes possible but also increases the danger of thermal runaway if the device is not adequately heatsunk.
TI Information - Selective Disclosure

43. On Board Charger: Error Amplifiers (I and V)
Measure output current
Compare to reference
Output error signal (power demand)
Measure output voltage
Diode ‘or’ errors – lowest error ‘wins’
Automatic CV / CI transition
This is the usual technique
Lowest error ‘wins’ and controls the output
Low side sense at 400Vout
High side sense at 12Vout is possible
TI Information – Selective Disclosure 43
The charger has to switch between CV and CI regulation like the typical characteristic shown at the bottom left. There are two error amplifiers, one comparing the output current against the constant current regulation setpoint and the other comparing the output voltage against the constant voltage regulation setpoint. If the system is in CI mode, the voltage error amplifier is saturated high because Vo is too low, if the system is in CV mode the current error amplifier saturates high because the current is below the regulation setpoint. The outputs from the two error amplifiers are diode ored and the lower of the two errors is fed into the controller. Note that the error amplifier in the UCC28951-Q1 is configured as a voltage follower and other than this it plays no part in the control loop. The on-board reference of the UCC28951-Q1 is not accurate enough for Li Ion battery charging in particular so an external LM4132-Q1 reference is used.
Output voltage control for temperature compensation of the battery voltage and for setting the float voltage in the lead acid battery charger is accomplished by a digital potentiometer controlled by an external microcontroller over an I2C bus
The TPL0401A-10-Q1 device shown here is a single-channel, digital potentiometer with 128 wiper positions.
TI Information - Selective Disclosure

44. On Board Charger: Input current sensing
Current Transformer in the input power rail senses input current
In this position, it senses the full bridge current
Senses any ‘shoot through’ events
QA and QB or QC and QD ON simultaneously
CS signal used for Peak Current Mode (PCM) control of PSFB
PCM gives cycle-by-cycle control of peak current in primary
Protection against transformer saturation
CS signal is used for regulation in both CV and CI modes
Regulation setpoint depends on whether the CV or CI error amplifier is in control
TI Information – Selective Disclosure 44
The PSFB in this design is controlled in Peak Current Mode (PCM). This means that the error signal from either the voltage or current error amplifier sets a current demand level for the inner PWM loop. We use a current transformer in the input power rail to sense the input current and the inner current loop compares the CS signal to the demand from the error amplifier to control the current in the usual way.
The input current is normally sensed by a Current Transformer located on the HV Bus as shown above. In this location, it senses the full bridge current and will also detect any shoot through currents in the bridge.
Peak Current Mode control allows the UCC28951-Q1 to provide cycle-by-cycle control of the peak current in the primary switches and prevents transformer saturation due to unbalanced primary currents. Of course we cannot use the primary current signal for CI regulation because the primary current is a function of the input voltage and there is no fixed relation between primary and secondary current. This is why we need the current error amplifier to set a demand level for the control loop and so regulate the output current.
TI Information - Selective Disclosure

45. Summary TI Information - Selective Disclosure
We have seen that the UCC28951-Q1 PSFB may be used in unidirectional DC/DC converters for charging high and low voltage batteries from high voltage sources like PFC outputs. This controller and topology can cover the power range from  several hundred Watts to several kW.  
The UCC28951-Q1 can play a big role in chargers, where the power level is relatively low. For charger applications some external intelligence needs to be added, current sensing, a small Micro for control monitoring, temperature compensation of output voltage etc. At the same time the UCC28951-Q1 can manage the power stage.
For higher power applications Micro controllers play the main role due to high power requirement of the OBC (<20kW for on board, kW for off board). However UCC28951-Q1 could also cover some of these applications – especially if Multiple-Phase or Master/Slave techniques were used.
TI Information – Selective Disclosure 45
We have seen that the UCC28951-Q1 PSFB may be used in unidirectional DC/DC converters for charging high and low voltage batteries from high voltage sources like PFC outputs. This controller and topology can cover the power range from  several hundred Watts to several kW.  
The UCC28951-Q1 can play a big role in chargers, where the power level is relatively low. For charger applications some external intelligence needs to be added, current sensing, a small Micro for control monitoring, temperature compensation of output voltage etc. At the same time the UCC28951-Q1 can manage the power stage.
For higher power applications Micro controllers play the main role due to high power requirement of the OBC (<20kW for on board, kW for off board). However UCC28951-Q1 could also cover some of these applications – especially if Multiple-Phase or Master/Slave techniques were used.
TI Information - Selective Disclosure

46. Summary TI Information - Selective Disclosure
MCU for system supervision
TI Information – Selective Disclosure 46
This is a reminder of the block diagram we have discussed. I’ll leave it on the screen for a few seconds more
TI Information - Selective Disclosure

47. Thank You
Colin Gillmor, HPC 47

48. TI Information - Selective Disclosure
This slide should be retained for the recording… Leave on screen for 5 seconds.
TI Information - Selective Disclosure