1. 1 Connectors, Cables, and Electromagnetic Compatibility (EMC) Chris Allen (callen@eecs.ku.edu) Course website URL people.eecs.ku.edu/~callen/713/EECS713.htm

2. 2 Connectors Purpose Transmit signals (& DC levels) across board boundaries while preserving signal fidelity Issues factors that can affect signal fidelity (T r, noise level) Crosstalk (inductive) Series inductance Electromagnetic interference Ground continuity Capacitance

3. 3 Connectors Consider the connection between two printed-circuit boards Assume the connector length, H, is electrically short i.e., H << ℓ, ℓ = v p T r Consider the return current path – both current paths X and Y share the same ground pin Part of the magnetic flux from signal X is enclosed in signal Y’s circuit path  The result is mutual inductance between X and Y, L XY mutual inductance  crosstalk Estimate of L XY includes contribution by signal and return current L XY has a component from the current overlap L XY has a component from the ground pin’s inductance

4. 4 Connectors To first order where a = distance from signal X to signal Y (inches) b = distance from signal Y to ground pin (inches) c = distance from signal X to ground pin (inches) D = diameter of connector pin (inches) H = pin length in connector (inches) L XY = mutual inductance between loops X & Y (nH) Overlap Ground pin

5. 5 Connectors Example Consider a standard connector with 100-mil pitch, 500-mil connector pin length, 10-mil pin diameter Find L XY for case when X, Y, and return path are in adjacent pins a = 100 mils b = 100 mils c = 200 mils H = 0.5” D = 0.010 ” 80% of L XY is from ground pin 20% from overlap Inductance from the ground pin is typically larger than the overlap term

6. 6 Connectors The induced signal, E, is Half of the crosstalk signal is launched in the forward direction, half in the reverse direction, therefore the induced voltage is L XY /(2RT r ) Consider GaAs gates ( T r = 150 ps ) with R = Z o = 50  f or L XY = 9.37 nH, the crosstalk will be 0.62 The coupled signal is 60% of the full signal swing mostly due to the shared ground pin Lesson learned – mutual inductance is mostly responsible for crosstalk generated by connectors

7. 7 Connectors How to reduce crosstalk in the connector? Increase T r  may adversely affect system performance Increase Z o and R Decrease L XY Now explore ways to reduce L XY Recall that 80% if L XY came from the ground pin term  changing the pin spacing has little effect To reduce L XY we can provide parallel ground pins

8. 8 Connectors After adding a ground pins, reduces L XY For this case the return current through the lower ground pin is reduced 50% the mutual inductance due to overlap similarly reduced. Likewise the addition of a second ground pin reduces the mutual inductance due to ground-pin inductance is reduced by 50%. However benefits from adding further ground-pin are reduced. Overlap Ground pin

9. 9 Connectors To reduce L XY further, separate X & Y with intervening ground pins For this case ground pins are added between the X & Y signal paths the ground-pin inductance becomes less significant than the overlap L XY is reduced by approximately 1/(1+N 2 ) For N = 5 Adding ground pins outside X & Y does little to reduce crosstalk Summary – use plenty of extra ground pins; separate signal pins as much as possible

10. 10 Series inductance Just as we saw with PC board traces, a disruption in the return current path results in an increased inductance While the return current path follows the path of least inductance, a portion may pass through GND1 or GND3 The result will be a larger current loop that introduces series inductance Longer risetime Potential crosstalk Radiation of energy – electromagnetic interference (EMI)

11. 11 Series inductance Guidelines for reducing series inductance Use extra ground pins in connectors provides low impedance return path Place all connectors with common elements close together reduces loop area Provide a low impedance ground on both boards reduces loop area

12. 12 Ground continuity beneath connector Ground pins in connector provide a return path for the signal current Placement of the ground pins and how it connects to the ground plane is important The large slot in the ground plane causes the return current to take a longer path

13. 13 Ground continuity beneath connector A preferred layout provides path for the return current

14. 14 Ground continuity beneath connector Should also consider reassigning the pins on the connector to avoid having all ground pins on the far side

15. 15 High-speed signal connectors Custom connectors have been designed for high-speed applications Characteristics – Internal ground structures Low-impedance ground path Deliberately increased pin capacitance to balance the pin inductance Amphenol NeXLev Mezzanine Connector Amphenol NeXLev Characteristics Ball grid array attach for high density surface mount Handles data rates up to 12 Gb/s High signal density – Up to 145 real single-ended signals per linear inch Approx 30 lb mating force for 300 I/O module

16. 16 High-speed signal connectors Custom connectors have been designed for high-speed applications Amphenol XCede Backplane Connector Designed for 25 Gb/s Performance 85 and 100 ohm impedance High signal density – Up to 84 differential signals per inch Routing out in two layers. First layer shown in blue, second layer shown in red. Double ground vias are spaced 1.56 mm apart providing a wide secondary routing channel.

17. 17 Flex circuits as connectors/cables Flexible circuits is a technology for assembling electronic circuits by mounting electronic devices on flexible plastic substrates, such as polyimide Can support controlled-impedance transmission lines (e.g., 50  ) in microstrip or stripline geometries Low observed crosstalk (< 2% in microstrip) Different types of flex circuits are available Single Layer Multiple Layer Double Layer Rigid Flex

18. 18 Flex circuits as connectors/cables Flexible circuits is a technology for assembling electronic circuits by

19. 19 Ribbon cables In digital systems, multiple signals are passed between boards For wide-bandwidth signals, controlled-impedance transmission lines are typically used At low frequencies, ribbon cables are often used Ribbon cables, like most other cable geometries, have limited frequency response due to the skin effect As a result, the risetime becomes degraded (longer T r ) and this becomes worse with cable length Crosstalk can also be significant As in the connector, this depends on placement of ground conductors for return the current path Can be reduced by introducing extra grounds G – S – G – G – S – G – G – S – G …

20. 20 Ribbon cables Alternative assignments of conductors within ribbon cable

21. 21 Ribbon cables To reduce the radiation from the cable, various techniques can be applied Twist differential wire pairs Shield the entire cable with conductive layer (e.g., foil) Add a choke Flat ribbon cable with periodically twisted adjacent wire pairs are commercially available

22. 22 Ribbon cables Shielded ribbon cable Continuous conductive shield contains the electric field and provides return current path for any stray currents Ensuring ground continuity at the connectors is essential to achieve the desired performance

23. 23 Ribbon cables Balanced and unbalanced lines A balanced line is a pair of similar conductors that have equal impedances along their length and have equal impedances to ground and to other circuits. Conductors in an unbalanced line have dissimilar conductors or do not have equal impedances to ground or other circuits Balanced and unbalanced circuits can be interconnected using a transformer called a balun Compared to unbalanced circuits, balanced circuits have better rejection of external noise BalancedUnbalanced

24. 24 Electromagnetic compatibility (EMC) EMC pertains to the ability of a device, equipment, or system to function satisfactorily in its electromagnetic (EM) environment without introducing intolerable EM disturbances to anything in that environment The two main aspects of EMC are function satisfactorily – EM susceptibility (EMS) without producing intolerable disturbances – EM interference (EMI) EMI standards are set by regulatory agencies Commercial products in the US Federal Communication Commission (FCC), Part 15 Military products in the US MIL-STD-461E EMS standards are set according to the application Regulated: medical devices, military Unregulated: commercial products (market driven)

25. 25 FCC regulations for digital devices For high-speed digital systems, EMS (susceptibility) is a measure of how well the system will perform its function in the presence of EM signals example, next to an AM or FM radio transmitter, near an airport with a powerful radar EMI for ‘digital devices’ is regulated by FCC for “An unintentional radiator (device or system) that generates and uses timing signals or pulses at a rate in excess of 9,000 pulses (cycles) per second and uses digital techniques; inclusive of telephone equipment that uses digital techniques or any device or system that generates and uses radio frequency energy for the purpose of performing data processing functions such as electronics computations, operations, transformations, recording, filing, sorting, storage, retrieval, or transfer.” f clk  9 kHz

26. 26 FCC regulations for digital devices The FCC divides digital devices into two classes Class A – marketed for use in commercial, industrial, or business environment Class B – marketed for use in the residential environment Personal computers fall under Class B Class B regulations are stricter than Class A FCC regulations have the force of law fines and/or jail for willful violation Two forms of emissions are regulated Conducted emissions Radiated emissions Conducted emissions pertain to currents passed through the unit’s AC power cord and connected to the power grid. The concern is enhanced radiation due to the large “antenna” represented by the AC distribution network

27. 27 FCC regulations for digital devices FCC regulation on conducted emissions are concerned with the frequency range from 450 kHz to 30 MHz Although the emission to be controlled is current, the limits are in volts A line impedance stabilization network (LISN) in-line with the AC power cord converts the interfering current to a measurable voltage

28. 28 FCC regulations for digital devices Line impedance stabilization network (LISN) Purpose – prevent noise external to test from contaminating measurement present constant impedance (in frequency and site to site) to the product between phase & ground and between neutral & ground I P = V P /50, I N = V N /50 50  represents input to spectrum analyzer 1 k  ( R 1 ) is to provide discharge path for C 1 when 50  is not present FCC limits for conducted emissions Class A digital devicesClass B digital devices 0.45 to 1.705 MHz, 1000  V or 60 dB  V0.45 to 30 MHz, 250  V or 48 dB  V 1.705 MHz to 50 MHz, 3000  V or 69.5 dB  V

29. 29 FCC regulations for digital devices Second form of emission is radiated emission – electric and magnetic fields Regulation requires measurement of E-field only Measured field strength in  V/m or dB  V/m From 30 MHz to 40 GHz Both V & H polarizations with respect to ground plane of test site Measured at a specific distance To be measured in open-field test site over a ground plane with tuned dipole Difficult to accomplish – so preliminary tests are made with Broadband antennas (biconical or log-periodic antennas) In a semi-anechoic chamber 100-kHz BW quasi-peak detector

30. 30 FCC regulations for digital devices FCC limits for radiated emissions from digital devices Class A @ 10 m distance 30 to 88 MHz: 39 dB  V/m 88 to 216 MHz: 43.5 dB  V/m 216 to 960 MHz: 46 dB  V/m above 960 MHz: 49.5 dB  V/m Class B @ 3 m distance 30 to 88 MHz: 40 dB  V/m 88 to 216 MHz: 43.5 dB  V/m 216 to 960 MHz: 46 dB  V/m above 960 MHz: 54 dB  V/m In the far field, field strength goes as 1/r so at 200 MHz, while both classes have limit at 43.5 dB  V/m (150  V/m) Class B limit is tighter by 10/3 or 3.33 due to the closer measurement distance

31. 31 FCC regulations for digital devices How difficult is this level to achieve? Example TTL low-power Schottky logic ( T r  6 ns, I  8 mA ) test circuit using coplanar strip transmission line This simple circuit violates both Class A and B limits

32. 32 Radiated emissions In high-speed digital circuits radiated emissions > conducted emissions Radiated emission come from currents on wires (PCB traces) Current type is key in determining radiated emission level Two current types: differential-mode current common-mode current Consider two parallel conductors each carrying current I 1, I 2 These currents can be decomposed into differential-mode current, I D common-mode current, I C

33. 33 Radiated emissions I D, differential-mode currents: equal amplitude, opposite direction Differential currents: desired currents on transmission line Common-mode currents: undesired I C not predicted in transmission line analysis Typically, I C << I D however I C produces larger radiated emissions than I D Why? From Faraday’s law that in the far field Predicted in transmission line analysis In far field, differential-mode fields cancel In far field, common-mode fields add

34. 34 Radiated emissions Example Consider a 1-m long ribbon cable, with a 50-mil pitch When carrying a signal with 30-MHz, 20-mA differential current it produces a radiated emission of 100  V/m @ 3 m To produce the same radiated emission level with common-mode current requires 8  A (30-MHz frequency @ 3 m distance) 20 mA / 8  A = 2500 or 68 dB of difference Conclusion: The radiation efficiency of common-mode currents is more than 1,000,000 times greater than that of differential-mode currents How are common-mode currents created? Structural asymmetries, large loop areas

35. 35 Radiated emissions How to quantify the radiated emissions Consider the conductor pair Assumptions Sinusoidal currents Conductor spacing ( S ) is small, S << First the radiation from differential-mode current At a point in the plane of the conductors at a distance d from the midpoint | E D | increases with current ( I D ), frequency ( f ), length ( L ), and separation ( S ) and decreases with distance ( d )

36. 36 Radiated emissions Radiation from differential-mode current Note the term L  S = loop area = A The parameter of interest is where K 1 = 1.316 x 10 -14 /d letting d = 3 m (Class B) makes K 1 = 4.39 x 10 -15 Thus |E D /I D | increases with frequency at +40 dB/decade

37. 37 Radiated emissions Radiation from differential-mode current Recall that for digital signals for f > f CLK, the signal power spectral density changes at –20 dB/decade for f > F knee, the signal power spectral density changes at –40 dB/decade

38. 38 Radiated emissions Radiation from differential-mode current Combining the spectral response of differential-mode current radiation with the digital signal spectrum yields the expected spectral characteristics radiated by a digital differential-mode current To reduce E D we can: Reduce I D (technology limited) Reduce F knee (increase T r ) Reduce the loop area, A From a practical perspective reducing the loop area, A, is most realistic

39. 39 Radiated emissions Now the radiation from common-mode current At a point in the plane of the conductors at a distance d from the midpoint | E C | increases with current ( I C ), frequency ( f ), length ( L ), and decreases with distance ( d ) Note: linear frequency, not f 2 no dependence on separation ( S ) numerical scale factor is 8 orders of magnitude larger Can simply into the form where K 2 = 1.257 x 10 -6 /d or for d = 3 m, K 2 = 4.19 x 10 -7 |E C /I C | increases with frequency at +20 dB/decade

40. 40 Radiated emissions Radiation from common-mode current Combining the spectral response of common-mode current radiation with the digital signal spectrum yields the expected spectral characteristics radiated by a digital common-mode current To reduce E C we can: Reduce I C Reduce F knee (increase T r ) Reduce the length, L

41. 41 Radiated emissions Other differences in the radiation from common-mode and differential- mode currents Radiated fields from differential-mode currents are largest in the plane containing the currents and have nulls in the plane of symmetry E D is smaller for reduced S Radiated fields from common-mode currents are uniformly large in all directions E C is independent of S

42. 42 Radiated emissions Returning now to the example f CLK = 10 MHz T r = 6 ns (TTL LS) L = 7” (178 mm) S = 165 mils (4 mm)  V = 4 V Z o = 400   I D = 10 mA F knee ~ 210 MHz  T r ~ 2.4 ns Spectral shape indicates radiation is from common- mode current

43. 43 Radiated emissions Returning now to the example f CLK = 10 MHz, T r = 6 ns, L = 7” (178 mm), S = 165 mils (4 mm)  V = 4 V, Z o = 400   I D = 10 mA Assume I C = 100  A f |E D max| |E C max|. 10 MHz f CLK 3  V/m (9 dB  V/m)746 mV/m (117 dB  V/m) 200 MHz F knee 1249  V/m (62 dB  V/m)14.9 V/m (143 dB  V/m )

44. 44 Reducing radiated emissions To reduce EMI we can Reduce the common-mode currents, I C  E C Reduce the loop area, A  E D Reduce F knee  E D and E C To reduce F knee (increase T r ) We can insert a low-pass filter in the signal path if this does not impair the circuit performance To reduce common-mode currents We can use a common-mode choke For currents where signal and return pass through the magnetic choke (e.g., 1, 2), the presence of the choke has no effect. The number of turns = 0 in this inductor. For currents where the signal passes through the choke but the return does not (e.g., 3, 4), the choke increases the path inductance. Number of turns = 1 in this inductor.

45. 45 Reducing radiated emissions To enhance the choke’s effectiveness we can use multiple windings around the choke We can reduce the loop area by using shielding Signal conductors surrounded by a good conducting shield provides a low- impedance return path for the signals If signal and return conductors are both surrounded by good conductor shields, the common-mode current will induce a return current in the shield Radiated emissions are thus reduced Breaks in the shield can cause significant radiated emissions especially at the cable ends where the shield and connector join However multiple conductors within a shield may experience crosstalk

46. 46 Reducing radiated emissions Common-mode chokes suppress E C but do not help with E D Recall that E D varies as a function of  We can take advantage of this feature in suppressing E D Consider a twist in a conductor pair for L << ℓ = T r /v p = v p /(2 F knee ) = knee /2 In the far field, the E D fields tend to cancel Similarly, coupling into an adjacent straight wire will also cancel  reduces crosstalk This technique, twisting wire pairs, does not help reduce radiation from common-mode currents For adjacent twisted pair lines, crosstalk also cancels if twisted in a like direction and at similar intervals