1. Measuring and Qualifying Upstream Signals
SCTE New Jersey Chapter
9/13/07

2. My Business Card Larry Jump Regional Sales Engineer
Sunrise Telecom Broadband
tech support

3. Questions and Comments are Welcome

4. Better understand how to make upstream signals and measurements
Purpose
Better understand how to make upstream signals and measurements
What are the signal impairments on the reverse path
Here is what we expect to accomplish with this presentation.

5. Agenda
16 QAM Advantages and Challenges
Measurements and Impairments
Case Study

6. Why 16-QAM? Voice Peer to Peer
Higher upstream data throughput required for:
Voice
Peer to Peer
Channel bandwidth, MHz
Symbol rate, ksym/sec
QPSK raw data rate, Mbps
QPSK nominal data rate, Mbps
16-QAM raw data rate, Mbps
16-QAM nominal data rate, Mbps
0.200
160
0.32
~0.3
0.64
~0.6
0.400
320
1.28
~1.2
0.800
640
2.56
~2.4
1.60
1,280
~2.3
5.12
~4.8
3.20
2,560
~4.6
10.24
~9.0
By going to 16 QAM, the upstream data rate goes from 5.12 MBs to MBs

7. Upstream 16 QAM Challenges
Once interference occurs in voice the data cannot be retransmitted.
Measurements are more difficult because the signals are bursty.
16 QAM looses 3 dB of headroom because the maximum modem output is 55 dBmV as opposed to 58 dBmV for QPSK.

8. More Upstream Challenges with 16 QAM
16 QAM is less robust than QPSK
Requires better SNR and MER
QAM means that the carrier is amplitude modulated and therefore more susceptible to amplitude based impairments such as:
Ingress
Micro-reflections
Compression

9. What needs to be done before launching 16 QAM?
Making 16-QAM work reliably requires attention to several details:
The cable modem termination system (CMTS) configuration must be optimized for 16-QAM
Its also important to configure the CMTS parameter called modulation profile correctly.
The entire cable system needs to be DOCSIS-compliant
Return and Forward Sweep!
Leakage kept to levels below that required by the FCC
If it leaks out, it leaks in!

10. Recommended Network Specifications
Part 76 of the FCC Rules
DOCSIS for upstream and downstream
NCTA Recommended Practices for upstream carriers
PacketCable reference architecture is available from cablelabs

11. Upstream Signal Measurements

12. Spectrum Analyzer Upstream Measurements
Upstream Carrier Levels
Spectrum Analysis
Constellation Measurements and Diagnosis
MER, BER, and Constellation Analysis
Upstream Linear Distortion Measurements

13. Upstream Level Measurement The First Step
Verify the upstream carrier amplitude at the input to the CMTS upstream port is within spec.
Usually 0 dBmV at the input, some systems may vary.
Can be measured using peak power on the preamble of the carrier
An average power measurement could also be made on a constant carrier injected at the correct level.
Be careful of mixed modulation profile measurements; remember there is a 3 dB difference between QPSK and 16 QAM
Measure total power at the input to the CMTS (<35dBmV, TP)
Measure the upstream digitally modulated carrier amplitude, and confirm that the level is within 2 dB or 3 dB of what is expected. A typical value is 0 dBmV, but check the manufacturer’s specifications and CMTS configuration for your particular setup. What you should look for here is a large departure from the expected value—a couple dB or so is within the measurement accuracy window of most spectrum analyzers and the calibration accuracy of how a CMTS sets operating levels. Be sure to check signals at other reverse frequencies. For instance, there may be set top box, telephony and other upstream carriers present. Make certain these are not overloading the CMTS input. DOCSIS specifies that the total RF input power to a CMTS upstream port is not to exceed +35 dBmV. (Note: If RF signals for other cable services such as telephony, impulse-pay-per-view, and so forth are present in the upstream spectrum, make sure that their operating levels have been set using a technique such as power per Hertz. This will help to ensure that non-cable modem signals are not present at levels that are substantially different from the cable modem digitally modulated carrier at the CMTS upstream input.)

14. Upstream Level Measurement
The generally accepted way of measuring an upstream TDMA bursty carrier is in the zero span mode. Notice that the peak power of the preamble of the DOCSIS carrier is measured.
Measurement made in the zero span mode
Peak power of the preamble

15. Spectrum Analysis -CNR -C/I

16. DOCSIS 1.1 Upstream RF Channel Transmission Characteristics

17. Upstream CNR
Check the upstream carrier-to-noise, carrier-to-ingress, and carrier-to-interference ratios
DOCSIS assumes a minimum of 25 dB for all three parameters
This is measured at the CMTS upstream port
Remember that we loose 3 dB of dynamic range with 16 QAM
CNR and SNR are different measurements!
The correct noise power bandwidth is equal to the symbol rate of the upstream carrier
Can use noise power markers
Next, measure the upstream carrier-to-noise, carrier-to-ingress and carrier-to-interference ratios. DOCSIS specifies a minimum of 25 dB for each of these parameters, regardless of modulation format. Note that this is an in-channel specification, and is to be measured at the CMTS upstream input.
These measurements are normally made by putting the analyzer in peak hold for several minutes.

18. CNR is a measurement performed on RF signals
CNR or SNR
CNR is a measurement performed on RF signals
Raw carrier power to raw noise power in the RF transport path only
Ideal for characterizing network impairments
SNR is a pre-modulation or post-detection measurement performed on baseband signals
Includes noise in original signal, transmitter or modulator, transport path, and receiver & demodulator
Ideal for characterizing end-to-end performance—the overall signal quality seen by the end user
Remember the specification is for CNR measured on a spectrum analyzer. A CMTS measures SNR and there is a difference.

19. CMTS Upstream SNR Measurement
Broadcom burst demodulator chips used in a CMTS provides an upstream SNR estimate.
Other factors may degrade CMTS-reported SNR, even when CNR is good including improper modulation profiles, bad timing errors, and poor headend combiner/splitter isolation. These of course would be system return path problems.
Impulse noise and certain other fast transient impairments generally will not show up in CMTS SNR estimate.
Outside of the range of 15~25 dB CNR, the SNR estimate will diverge numerically from measured CNR value by as much as 4 dB or more.
CMTS-reported SNR will always be less than—or at best equal to—CNR, but should never be better than CNR.
The purpose of this discussion is do not rely on CMTS SNR estimates.

20. Upstream Spectrum Analysis
Make sure noise floor of system is being displayed 10 db out of the spectrum analyzer noise floor
Use peak hold to capture transients
Use Averaging to capture CPD

21. Good CNR and C/I
3.2MHz
After measuring the upstream digitally modulated carrier’s amplitude using the zero span method, switch the spectrum analyzer back to frequency domain mode (frequency vs. amplitude). Measure the amplitude of the system noise floor (be sure that you’re measuring the system’s noise floor and not the spectrum analyzer’s noise floor, AND make certain that when you measure the noise floor amplitude you correct the measurement to a noise power bandwidth equal to the digitally modulated carrier’s symbol rate). The difference between the digitally modulated carrier’s amplitude and the corrected noise floor amplitude is the carrier-to-noise ratio. Perform amplitude measurements of ingress and other interference as required. Here, too, the difference between the digitally modulated carrier’s amplitude and that of the interference is the carrier-to-interference ratio.

22. Effects of Over-Driving a Laser
The return path of a DOCSIS network is specified as 5-42 MHz (5-65 MHz Euro-DOCSIS) and RF amplifiers typically transport up to 42 MHz due to the diplex filters.
Return path optical transmitters transport frequencies up to 200 MHz. For this reason it is important to look at the return path in the headend all the way to 200 MHz when using a spectrum analyzer. This will enable the user to observe if the laser is operating in a non-linear (or clipping) mode.
The presence of frequencies above 42 MHz and a mirror image of the “hump” and DOCSIS channel on the right hand side is a clear indication that the laser is operating in a non-linear mode and creating distortion components.
When distortion components of this nature are present in an optical component, one can be certain that the laser is also clipping signals, causing lost symbols and thus dropped VoIP frames.
This will provide a clear indication of a poor VoIP channel and dissatisfied VoIP customers.
Notes
___________________________________________________________

23. CNR using a Noise Marker
A quick way to measure the downstream digitally modulated carrier’s CNR is with a spectrum analyzer’s marker noise function (1 Hz noise power bandwidth).
The marker automatically extrapolates the power of the carrier into the power contained in 1 Hertz.
First measure the carrier amplitude, then the system noise floor. The difference between the two values is the approximate CNR—within a dB or so. Bandwidth and other corrections are not necessary. Be sure that system noise, and not analyzer noise is being measured! Temporarily disconnect the analyzer’s RF input. The observed noise floor amplitude should drop at least 10 dB. If it doesn’t, you’re measuring analyzer noise, not system noise.

24. Upstream Carrier-to-Interference
After measuring the upstream digitally modulated carrier’s amplitude using the zero span method, switch the spectrum analyzer back to frequency domain mode (frequency vs. amplitude). Measure the amplitude of the system noise floor (be sure that you’re measuring the system’s noise floor and not the spectrum analyzer’s noise floor, AND make certain that when you measure the noise floor amplitude you correct the measurement to a noise power bandwidth equal to the digitally modulated carrier’s symbol rate). The difference between the digitally modulated carrier’s amplitude and the corrected noise floor amplitude is the carrier-to-noise ratio. Perform amplitude measurements of ingress and other interference as required. Here, too, the difference between the digitally modulated carrier’s amplitude and that of the interference is the carrier-to-interference ratio.
The upper left photo shows a relatively clean upstream, with a carrier-to-noise ratio of ~40 dB, and carrier-to-interference ratio >30 dB. The top right photo shows a severely impaired upstream with excessive ingress and impulse noise. The C/I ratio is nominally 10~15 dB, and a substantial amount of packet loss was observed. The lower left photo shows impulse noise as well as in-channel interference apparently higher than the digitally modulated carrier. The lower right photo shows high levels of impulse noise, which caused significant packet loss.

25. Constellation Analysis
Patterns in the Constellation

26. MER, A Better Measurement
A better parameter than SNR is modulation error ratio (MER) or error vector magnitude (EVM)
MER takes into account:
CNR
Phase Noise (jitter of phase of QAM modulator’s carrier)
Intermod Distortions
Compression of Lasers and Amplifiers
Frequency Response
THE SUM OF ALL EVILS
MER is a single figure of merit for the quality of an RF QAM modulated signal.
MER and EVM are the same thing. MER is expressed in dB; EVM is expressed in %.
A direct measurement of the digital signals modulation quality
Can be directly linked to BER

27. 1011 Vectors and 16 QAM Q 90° I 180° I 0° Q 270° 11 1011 1111 10 1010
1110
I 180°
I 0° 00 01 10 11 01
This slide looks at a 16 QAM carrier. That means that each symbol is 4 bits (2 to the 4th = 16) In this slide if we wanted to transmit the bit stream 1010, we would modulate the I carrier with .35 volts and the Q carrier with .35 volts. This produces a resultant vector where that symbol is mapped in the constellation. The symbols are mapped so that if an error occurs in the vector, the resultant 4 bit word will only off by one bit.
We are also transmitting the digital bit stream using analog transmission techniques.
At the receive end the 2 carriers are demodulated and vectored to produce the original bit stream.
0001
0101 00
0000
0100
Q 270°

28. 1011 Vectors and 16 QAM Q 90° I 180° I 0° Q 270° 11 1011 1111 10 1010
1110
I 180°
I 0° 00 01 10 11 01
In this example ingress from the coaxial plant has changed the levels of the I and Q carriers before they arrive at the digital receiver. As a result of this the new vector is not on the target anymore. There is then also a vector difference, shown in blue, between the target and the where the symbol actually falls. The difference between the target and the new resultant vector is MER or Modulation Error Ratio.
0001
0101 00
0000
0100
Q 270°

29. Modulation Error Ratio
RMS error magnitude
average symbol magnitude
10 log
MER is defined as follows:
MER is expressed in dB.
RMS Error Magnitude
Ideal Symbol
This slide shows a magnified view of one quarter of a constellation. It is called a ZOOM view. This is only the upper right hand quadrant of the constellation shown here giving us a view of the upper right 16 data points out of the 64 possible data points. MER is definded as the average symbol amplitude (approximately the distance between the origin or center of the graph and the decision point for the symbol) divided into the RMS error magnitude (in other words, how much the actual phase and amplitude of the signal missed the target for that particular symbol).
As the signal is degraded and the plotted points land farther and farther away from the center of the decision window for each symbol, the MER will become lower and lower, but the QAM will remain locked until the failure point.
Average Symbol Magnitude

30. Modulation Error Ratio
Minimum recommended downstream MER (includes 3 to 4 dB of headroom for reliable operation)
256-QAM: 31 dB
QPSK: 13 dB
16 QAM: 20 dB
Graphic courtesy of Sunrise Telecom

31. Introduction to BER
Bit Error Rate (BER) is an important concept to understand in any digital transmission system since it is a major indicator of the quality of the digital system.
As data is transmitted some of the bits may not be reproduced at the receiver correctly. The more bits that are incorrect, the more the signal will be affected.
BER is a ratio of incorrect bits to the total number of bits measured.
Its important to know what portion of the bits are in error so you can determine how much margin the system has before failure.

32. What is BER? Sent Bits 1101101101 Received Bits 1100101101 1
BER is defined as the ratio of the number of wrong bits over the number of total bits.
BER is measured by sending a known string of bits and then counting the errored bits vs. the total number of bits sent.
This is technically an out of service measurement.
Sent Bits
Received Bits
error
# of Wrong Bits 1
BER = = =
0.1
# of Total Bits 10

33. What is BER? Lower and Better BER
BER is normally displayed in Scientific Notation.
The more negative the exponent, the better the BER.
Better than 1.0E-6 is needed after the FEC for the system to operate.
Lower and Better BER

34. Forward Error Correction
In every MPEG digital receiver, there is a FEC decoder that can actually repair damaged data.
Forward error correction (FEC) is a digital transmission system that sends redundant information along with the payload, so that the receiver can repair the damaged data and eliminate the need to retransmit.
It only works if the BER is higher than 1E-6 before the FEC decoder
Reed Solomon can correct bit errors down to approximately 1 bit for every 1000 (10-6). If you have bit errors at 10-6, you need to start looking for your problems.

35. Pre and Post FEC BER
BER performance you need to know both the pre and post FEC bit error rate.
The FEC decoder needs a BER of better than 1 E-6 in order to operate.
Post FEC Bit errors are not acceptable. To get an accurate idea of the
You should look at both the Pre and Post FEC BER to determine if the FEC is working to correct errors and if so how hard.
A Pre BER measurement of 1E-6 is required at the input so that no Post FEC bit errors are present.
Pre FEC BER
FEC Decoder
Post FEC BER

36. Noise and Intermittents
Errors caused by noise or intermittent causes can have the same BER, but very different effects.
Errors that are spread out are due to noise problems
Errors that are grouped are due to intermittent problems such as ingress or loose connectors.
Spaced Errors
Burst Errors
Notice that it would make sense that the burst errors are much more difficult to correct because there are many errors in a row. This is why interleaving is used.
This Example Shows the Same Error Rate But the Burst Errors are More Difficult to Correct

37. Return Path Constellation Analysis

38. 16 QAM Upstream Analysis Return Path Verification, Test
& Troubleshooting

39. CPD and Noise
Notice the scattered symbols and that the MER has degraded to 30 from a 35

40. Laser Clipping
Notice the corners being pulled in toward the center

41. Noise
Constellation shows noise and an MER of 21, just out of the danger zone. No BER present.

42. Ingress
Notice the widely scattered symbols

43. A Good 16 QAM Constellation
Zero Bit Errors
The goal is to have no bit errors at the headend

44. Adaptive Equalizers Corrects for Frequency Response imperfections
Corrects for Group Delay
Show impedance mismatches

45. Adaptive Equalizers
Most digital receivers have an adaptive equalizer. Adaptive Equalizers optimize the received signals before they are demodulated. It attempts to maximize the MER.
Adaptive Equalizers in a digital receiver perform 3 functions
1.They correct for amplitude response problems
2.They correct for group delay problems
3.They ring at the symbol rate and only allow 1 symbol to enter the DAC at a time. If there are reflections due to impedance mismatches then there will be inter-symbol interference.
This graphic shows equalizer stress, in-channel frequency response and group delay ripple (arrow) for a 256-QAM digitally modulated carrier. The worst-case peak-to-peak group delay ripple shown here is about 40 nanoseconds, well within the DOCSIS 75 ns assumed maximum.
Adaptive equalizers cannot compensate for:
Noise
I/Q phase or gain impairments
Compression

46. Microreflections Micro-reflections are impedance mismatches
In the real world of cable networks, 75 Ω impedance is at best considered nominal
Micro-reflections cause group delay and frequency response problems.
Impedance mismatches are everywhere: connectors, amplifiers inputs and outputs, passive device inputs and outputs, and even the cable itself
Upstream cable attenuation is lower than downstream cable attenuation, so upstream micro-reflections tend to be worse.
Anywhere an impedance mismatch exists, some of the incident energy is reflected back toward the source

47. Micro-reflections
Higher orders of modulation are affected by micro-reflections to a much greater degree so 16 QAM is affected more than QPSK
Upstream micro-reflections and group delay are minimized by using adaptive equalizers. This feature is available in DOCSIS 1.1 and 2.0 CMTSs , but not 1.0.

48. Causes: Microreflections Damaged or missing end-of-line terminators
Damaged or missing chassis terminators on directional coupler, splitter, or multiple-output amplifier unused ports
Loose center conductor seizure screws
Unused tap ports not terminated—this is especially critical on low value taps
Unused drop passive ports not terminated
Use of so-called self-terminating taps at feeder ends-of-line

49. Causes (cont’d): Microreflections
Kinked or damaged cable (includes cracked cable, which causes a reflection and ingress)
Defective or damaged actives or passives (water-damaged, water-filled, cold solder joint, corrosion, loose circuit board screws, etc.)
Cable-ready TVs and VCRs connected directly to the drop (return loss on most cable-ready devices is poor)
Some traps and filters have been found to have poor return loss in the upstream, especially those used for data-only service

50. -10 dBc @ <=0.5 µsec -20 dBc @ <=1.0 µsec -30 dBc @ >1.0 µsec
Microreflections
-10 <=0.5 µsec
-20 <=1.0 µsec
-30 >1.0 µsec
Microreflections—also called reflections or echoes—are caused by impedance mismatches
In the real world of cable networks, impedance can at best be considered nominal
Impedance mismatches are everywhere: connectors, amplifiers inputs and outputs, passive device inputs and outputs, and even the cable itself
Upstream cable attenuation is lower than downstream cable attenuation, so upstream microreflections tend to be worse
Anywhere an impedance mismatch exists, some of the incident energy is reflected back toward the source
The reflected and incident energy interact to produce standing waves, which manifest themselves as the “standing wave” amplitude ripple one sometimes sees in sweep receiver displays
16-QAM is affected by microreflections to a much greater degree than QPSK is
Microreflections and group delay may be compensated for using adaptive equalization, a feature available in DOCSIS 1.1 and 2.0 cable modems
Adaptive equalization is not supported by most DOCSIS 1.0 modems
Causes
Damaged or missing end-of-line terminators
Damaged or missing chassis terminators on directional coupler, splitter, or multiple-output amplifier unused ports
Loose center conductor seizure screws
Unused tap ports not terminated—this is especially critical on low value taps
Unused drop passive ports not terminated
Use of so-called self-terminating taps at feeder ends-of-line
Kinked or damaged cable (includes cracked cable, which causes a reflection and ingress)
Defective or damaged actives or passives (water-damaged, water-filled, cold solder joint, corrosion, loose circuit board screws, etc.)
Cable-ready TVs and VCRs connected directly to the drop (return loss on most cable-ready devices is poor)
Some traps and filters have been found to have poor return loss in the upstream, especially those used for data-only service

51. Microreflections
Here’s an example: An approx. -33 dBc echo at just over 1 µsec
This echo meets the DOCSIS upstream -30 dBc at >1.0 µsec parameter however this is sufficient to cause some amplitude and group delay ripple

52. Amplitude Ripple ( Frequency Response)
“Amplitude ripple” is also known as frequency response.

53. Amplitude Ripple An in-service spectrum analyzer measurement
A spectrum analyzer may be used to characterize a upstream digitally modulated carrier’s in-channel frequency response, essentially using the noise-like signal to measure itself. This method eliminates the need to take the digitally modulated carrier out of service, and does not require a sweep signal.
An in-service spectrum analyzer measurement

54. Group delay is worse near band edges and diplex filter roll-off areas
Different travel through the same medium at different speeds. This is Group Delay
Group delay is defined in units of time, typically nanoseconds (ns) over frequency. In other words how much GD per each MHz.
In a system, network or component with no group delay, all frequencies are transmitted through the system, network or component with equal time delay
Frequency response problems in a CATV network will cause group delay problems
Group delay is worse near band edges and diplex filter roll-off areas
In simplified terms, when there is no group delay in a system, network or component, all frequencies within a defined bandwidth will take the same amount of time to traverse that system, network or component. When group delay does exist, signals at some frequencies will arrive at slightly different times than signals at other frequencies will.
Group delay in a CATV network is usually related to frequency response problems, and is most often a problem near band edges and diplex filter rolloff regions.

55. Group delay may still be a problem when the frequency response is flat
Upstream frequency
Keep the 16-QAM digitally modulated carrier well away from diplex filter roll-off areas (typically above about 35~38 MHz), where group delay can be a major problem
Choose an operating frequency that will minimize the likelihood of group delay
Frequencies in the 20~35 MHz range generally work well
Group delay may still be a problem when the frequency response is flat

56. Group Delay
“Amplitude ripple” is also known as frequency response.

57. Group Delay Measurement

58. Some things to check out!
Before adding a 16 QAM carrier the following should be checked
Compression of the return laser due to added carrier or a carrier with added bandwidth
MER and BER over a period of time
Group Delay of a new carrier
MER and BER of the new carrier.
Amplitude Ripple
Microreflections

59. Upstream Spectrum Display Showing Compression
This slides shows second order products being generated by the laser clipping above the return diplexer frequency. This picture is at the headed with a 16 QAM carrier being generated in the field. The second order products are not the result of past the node because the diplexers would attenuate them above the cut off frequency.
Some things to watch for:
Look below 5 MHz! The bandpass of the return path is 0-42 MHz not 5-42 MHz. The power below 5 MHz will compress your lasers as well.
Look above 42 MHz, even up to 200 MHz. If diplexers do not have good isolation between the ports, the forward carriers will be bleed into the return path.
3.If the laser clips, it will also have the tendency to spread the bottom of your digital carrier out.
4.Look for CPD and ingress in the spectrum mode

60. Statistics Mode
The Statistics mode shows MER, and Pre and Post BER over a period of time. This mode would capture problems that would not show up in instantaneous measurements.

61. Group Delay Measurement
The vertical axis is time and the horizontal axis is frequency.
The DOCSIS specification for group delay is 200 nSec over a 1 MHz bandwidth, however 100 nSec is becoming the more accepted standard.

62. Frequency Response of an Upstream Carrier
The DOCSIS specification is for 3 dB P/V

63. A Case Study

64. Upstream Spectrum
The upstream spectrum here shows some ingress at the lower frequencies and a CNR of about 35 dB. The green marker is at the diplexer cut off frequency. Notice there is very little second order above that frequency.
16 QAM did not work here even though specs for CNR, and C/I are well above the specification of 30

65. Noise
Constellation shows noise and an MER of 21, just out of the danger zone. No BER present.

66. Oops!
The arrow shows a microreflection at 2.5uSecs at -23 dBc. The DOCSIS spec is 30.
This reflection is high enough to cause amplitude ripple and severe group delay

67. Amplitude Ripple D = 492(VP/F) = 492(87%/≈.4MHz) ≈ 1100 feet
the amplitude ripple peaks are about 0.4 MHz apart. Using the formula D = 492(VP/F)—where D is distance in feet, VP is the cable’s velocity of propagation (87%), and F is the frequency spacing in MHz of the standing wave ripples (0.4 MHz)—the estimated distance to the microreflection point is just under 1,100 feet.
D = 492(VP/F) = 492(87%/≈.4MHz) ≈ 1100 feet

68. Group Delay
Peak to Valley Group Delay ≈ 270 nSeconds

69. Bit Errors
This is statistic mode shot of the injected 23 MHz carrier showing the Pre FEC bit errors caused by the laser clipping

70. Moral of the Story?
CNR and Distortion measurements from a spectrum analyzer are great but, don’t tell the whole story.
Other digital measurements are advised using a vector analyzer to ensure 16 QAM reliability
MER and BER
Group Delay and other Equalizer measurements
Constellation
Statistic Measurement

71. Measurement Summary Check for laser clipping Measure over time
Measure for frequency response of the carrier
Measure group delay of the carrier
Measure MER and BER of upstream carrier
Can be accomplished by inserting a 16 QAM carrier at the EOL and using a digital analyzer in the headend.

72. 16-QAM Pre-Launch Checklist
CMTS modulation profile optimized for 16-QAM
Vector Analysis, not just spectrum analysis
Entire cable network—headend, distribution network and subscriber drops—DOCSIS-compliant
Select upstream frequency that avoids diplex filter roll-off area
Forward and reverse properly aligned
Signal leakage and ingress management
Good installation practices

73. Ron Hranac wrote the book
References
Ron Hranac wrote the book
Hranac R., “CNR versus SNR” March 2003 Communications Technology
Hranac R., “Spectrum analyzer CNR versus CMTS SNR” September 2003 Communications Technology
Hranac R., “16 QAM Plant Preparation”
Hranac R., “Deploying VOIP on the Outside Plant”
Hranac R., “Linear Distortions,” Last 2 issues of CT Magazine

74. References
“RF Impairments in the Return Path and their impact on DOCSIS performance”, by Jack Moran, Motorola
National Cable Television Association’s Recommended Practices for Measurements on Cable Television Systems, 2nd Edition, October 1997 “Supplement on Upstream Transport Issues.”
“Broadband Return Systems for HFC Cable TV Networks”, by Donald Raskin and Dean Stoneback
“Return Path Level Selection, Set Up, and Alignment Procedure”, Motorola 1997
“Modern Cable Television Technology”, by Walter Cicora, James Farmer and David Large

75. More References
“Mystified by Return Path Activation?” Get your Upstream Fiber Links Aligned, by Ron Hranac, Communications Technology, March 2000
“Seek Balance in All Things” A Look at Unity Gain in the Upstream Coax Plant, by Ron Hranac, Communications Technology, June 2000
“A Primer on Common Path Distortion”, by Nick Romanick, Communications Technology, April 2001

76. Thank You!